Apparatus for transmitting broadcast signals, apparatus for receiving broadcast signals, method for transmitting broadcast signals and method for receiving broadcast signals

ABSTRACT

The present invention provides an apparatus of transmitting broadcast signals. The apparatus includes, an encoder for encoding service data, a bit interleaver for bit interleaving the encoded service data, a mapper for mapping the bit interleaved service data into a plurality of OFDM (Orthogonal Frequency Division Multiplex) symbols to build at least one signal frame, an OFDM modulator for modulating data in the built at least one signal frame by an OFDM scheme and a transmitter for transmitting the broadcast signals having the modulated data.

This application claims the benefit of U.S. Provisional Patent Application Nos. 61/942,581 filed on Feb. 20, 2014, 61/942,584 filed on Feb. 20, 2014, 61/942,586 filed on Feb. 20, 2014, 61/950,196 filed on Mar. 9, 2014, 61/968,329 filed on Mar. 20, 2014, 62/002,154 filed on May 22, 2014 which is hereby incorporated by reference as if fully set forth herein.

BACKGROUND OF THE INVENTION Field of the Invention

The present invention relates to an apparatus for transmitting broadcast signals, an apparatus for receiving broadcast signals and methods for transmitting and receiving broadcast signals.

Discussion of the Related Art

As analog broadcast signal transmission comes to an end, various technologies for transmitting/receiving digital broadcast signals are being developed. A digital broadcast signal may include a larger amount of video/audio data than an analog broadcast signal and further include various types of additional data in addition to the video/audio data.

That is, a digital broadcast system can provide HD (high definition) images, multi-channel audio and various additional services. However, data transmission efficiency for transmission of large amounts of data, robustness of transmission/reception networks and network flexibility in consideration of mobile reception equipment need to be improved for digital broadcast.

SUMMARY OF THE INVENTION

Accordingly, the present invention is directed to an apparatus for transmitting broadcast signals and an apparatus for receiving broadcast signals for future broadcast services and methods for transmitting and receiving broadcast signals for future broadcast services.

An object of the present invention is to provide an apparatus and method for transmitting broadcast signals to multiplex data of a broadcast transmission/reception system providing two or more different broadcast services in a time domain and transmit the multiplexed data through the same RF signal bandwidth and an apparatus and method for receiving broadcast signals corresponding thereto.

Another object of the present invention is to provide an apparatus for transmitting broadcast signals, an apparatus for receiving broadcast signals and methods for transmitting and receiving broadcast signals to classify data corresponding to services by components, transmit data corresponding to each component as a data pipe, receive and process the data

Still another object of the present invention is to provide an apparatus for transmitting broadcast signals, an apparatus for receiving broadcast signals and methods for transmitting and receiving broadcast signals to signal signaling information necessary to provide broadcast signals.

To achieve the object and other advantages and in accordance with the purpose of the invention, as embodied and broadly described herein, the present invention provides a method of transmitting broadcast signals. The method of transmitting broadcast signals includes encoding service data, bit interleaving the encoded service data, mapping the bit interleaved service data into a plurality of OFDM (Orthogonal Frequency Division Multiplex) symbols to build at least one signal frame, modulating data in the built at least one signal frame by an OFDM scheme and transmitting the broadcast signals having the modulated data.

The present invention can achieve transmission flexibility by transmitting various broadcast services through the same RF signal bandwidth.

The present invention can improve data transmission efficiency and increase robustness of transmission/reception of broadcast signals using a MIMO system.

According to the present invention, it is possible to provide broadcast signal transmission and reception methods and apparatus capable of receiving digital broadcast signals without error even with mobile reception equipment or in an indoor environment.

BRIEF DESCRIPTION OF THE DRAWINGS

The accompanying drawings, which are included to provide a further understanding of the invention and are incorporated in and constitute a part of this application, illustrate embodiment(s) of the invention and together with the description serve to explain the principle of the invention. In the drawings:

FIG. 1 illustrates a structure of an apparatus for transmitting broadcast signals for future broadcast services according to an embodiment of the present invention.

FIG. 2 illustrates an input formatting block according to one embodiment of the present invention.

FIG. 3 illustrates an input formatting block according to another embodiment of the present invention.

FIG. 4 illustrates an input formatting block according to another embodiment of the present invention.

FIG. 5 illustrates a BICM block according to an embodiment of the present invention.

FIG. 6 illustrates a BICM block according to another embodiment of the present invention.

FIG. 7 illustrates a frame building block according to one embodiment of the present invention.

FIG. 8 illustrates an OFDM generation block according to an embodiment of the present invention.

FIG. 9 illustrates a structure of an apparatus for receiving broadcast signals for future broadcast services according to an embodiment of the present invention.

FIG. 10 illustrates a frame structure according to an embodiment of the present invention.

FIG. 11 illustrates a signaling hierarchy structure of the frame according to an embodiment of the present invention.

FIG. 12 illustrates preamble signaling data according to an embodiment of the present invention.

FIG. 13 illustrates PLS1 data according to an embodiment of the present invention.

FIG. 14 illustrates PLS2 data according to an embodiment of the present invention.

FIG. 15 illustrates PLS2 data according to another embodiment of the present invention.

FIG. 16 illustrates a logical structure of a frame according to an embodiment of the present invention.

FIG. 17 illustrates PLS mapping according to an embodiment of the present invention.

FIG. 18 illustrates EAC mapping according to an embodiment of the present invention.

FIG. 19 illustrates FIC mapping according to an embodiment of the present invention.

FIG. 20 illustrates a type of DP according to an embodiment of the present invention.

FIG. 21 illustrates DP mapping according to an embodiment of the present invention.

FIG. 22 illustrates an FEC structure according to an embodiment of the present invention.

FIG. 23 illustrates a bit interleaving according to an embodiment of the present invention.

FIG. 24 illustrates a cell-word demultiplexing according to an embodiment of the present invention.

FIG. 25 illustrates a time interleaving according to an embodiment of the present invention.

FIG. 26 illustrates the basic operation of a twisted row-column block interleaver according to an embodiment of the present invention.

FIG. 27 illustrates an operation of a twisted row-column block interleaver according to another embodiment of the present invention.

FIG. 28 illustrates a diagonal-wise reading pattern of a twisted row-column block interleaver according to an embodiment of the present invention.

FIG. 29 illustrates interleaved XFECBLOCKs from each interleaving array according to an embodiment of the present invention.

FIG. 30 shows a parity check matrix of a QC-IRA (quasi-cyclic irregular repeat accumulate) LDPC code.

FIG. 31 shows a process of encoding the QC-IRA LDPC code according to an embodiment of the present invention.

FIG. 32 illustrates a parity check matrix permutation process according to an embodiment of the present invention.

FIGS. 33, 34 and 35 illustrate a table showing addresses of parity check matrix according to an embodiment of the present invention.

FIGS. 36 and 37 illustrate a table showing addresses of parity check matrix according to another embodiment of the present invention.

FIG. 38 illustrates a method for sequentially encoding the QC-IRA LDPC code according to an embodiment of the present invention.

FIG. 39 illustrates an LDPC decoder according to an embodiment of the present invention.

FIG. 40, FIG. 41, FIG. 42 and FIG. 43 illustrate one of the embodiments of the H1 matrix according to a code rate of 7/15.

FIG. 44 and FIG. 45 illustrate one of the embodiments of the H2 matrix according to a code rate of 7/15.

FIG. 46 illustrates one of the embodiments of the degree distribution table according to a code rate of 7/15.

FIG. 47, FIG. 48 and FIG. 49 illustrate one of the embodiments of the H1 matrix according to a code rate of 8/15.

FIGS. 50 and 51 illustrate one of the embodiments of the H2 matrix according to a code rate of 8/15.

FIG. 52 illustrates one of the embodiments of the degree distribution table according to a code rate of 8/15.

FIG. 53, FIG. 54 and FIG. 55 illustrate one of the embodiments of the H1 matrix according to a code rate of 11/15.

FIG. 56 and FIG. 57 illustrate one of the embodiments of the H2 matrix according to a code rate of 11/15.

FIG. 58 illustrates one of the embodiments of the degree distribution table according to a code rate of 11/15.

FIG. 59, FIG. 60, FIG. 61 and FIG. 62 illustrate one of the embodiments of the H1 matrix according to a code rate of 12/15.

FIG. 63, FIG. 64, and FIG. 65 illustrate one of the embodiments of the H2 matrix according to a code rate of 12/15.

FIG. 66 illustrates one of the embodiments of the degree distribution table according to a code rate of 12/15.

FIG. 67 and FIG. 68 are a table showing addresses of parity check matrix according to an embodiment of the present invention.

FIG. 69 is a table showing addresses of parity check matrix according to another embodiment of the present invention.

FIG. 70 and FIG. 71 is a table showing addresses of parity check matrix according to another embodiment of the present invention.

FIG. 72 is a table showing addresses of parity check matrix according to another embodiment of the present invention.

FIG. 73 and FIG. 74 is a table showing addresses of parity check matrix according to another embodiment of the present invention.

FIG. 75 is a table showing addresses of parity check matrix according to another embodiment of the present invention.

FIG. 76 and FIG. 77 is a table showing addresses of parity check matrix according to another embodiment of the present invention.

FIG. 78 is a table showing addresses of parity check matrix according to another embodiment of the present invention.

FIG. 79 is a table showing addresses of parity check matrix according to another embodiment of the present invention.

FIG. 80 is a table showing addresses of parity check matrix according to another embodiment of the present invention.

FIG. 81 is a table showing addresses of parity check matrix according to another embodiment of the present invention.

FIG. 82 is a table showing addresses of parity check matrix according to another embodiment of the present invention.

FIG. 83 illustrates a constellation mapper according to one embodiment of the present invention.

FIG. 84 describes a process of mapping IQ-balanced/IQ-symmetric non-uniform constellations according to one embodiment of the present invention.

FIG. 85 shows constellations of 64 NUCs at the SNR of 18 dB using the method of IQ-balanced non-uniform constellation mapping according to one embodiment of the present invention.

FIG. 86 shows a constellation of 64 NUCs for 10/15 code rate based on the IQ-balanced non-uniform constellation mapping method according to one embodiment of the present invention.

FIG. 87 shows a constellation of 64 NUCs for 10/15 code rate based on the IQ-symmetric non-uniform constellation mapping method according to one embodiment of the present invention.

FIG. 88 shows a constellation of 256 NUCs for 10/15 code rate based on the IQ-balanced non-uniform constellation mapping method according to one embodiment of the present invention.

FIG. 89 shows a part of the coordinates of the constellations of 256 NUCs for 10/15 code rate based on the IQ-balanced non-uniform constellation mapping method according to one embodiment of the present invention.

FIG. 90 shows another part of the coordinates of the constellations of 256 NUCs for 10/15 code rate based on the IQ-balanced non-uniform constellation mapping method according to one embodiment of the present invention.

FIG. 91 shows the other part of the coordinates of the constellations of 256 NUCs for 10/15 code rate based on the IQ-balanced non-uniform constellation mapping method according to one embodiment of the present invention.

FIG. 92 shows a constellation of 256 NUCs for 10/15 code rate based on the IQ-symmetric non-uniform constellation mapping method according to one embodiment of the present invention.

FIG. 93 shows a part of the coordinates of the constellations of 256 NUCs for 10/15 code rate based on the IQ-symmetric non-uniform constellation mapping method according to one embodiment of the present invention.

FIG. 94 shows another part of the coordinates of the constellations of 256 NUCs for 10/15 code rate based on the IQ-symmetric non-uniform constellation mapping method according to one embodiment of the present invention.

FIG. 95 shows the other part of the coordinates of the constellations of 256 NUCs for 10/15 code rate based on the IQ-symmetric non-uniform constellation mapping method according to one embodiment of the present invention.

FIG. 96 is a block diagram illustrating a bit interleaver according to an embodiment of the present invention.

FIG. 97 illustrates an inner group interleaving process for NUQ-256 according to an embodiment of the present invention.

FIG. 98 illustrates a write operation of inner group interleaving for NUC-256 according to an embodiment of the present invention.

FIG. 99 illustrates a read operation of inner group interleaving for NUC-256 according to an embodiment of the present invention.

FIG. 100 illustrates a remaining QC block operation according to an embodiment of the present invention.

FIG. 101 is a block diagram illustrating a demapping & decoding module 9020 according to an embodiment of the present invention.

FIG. 102 is a block diagram illustrating bit deinterleaving according to another embodiment of the present invention.

FIG. 103 is a table illustrating inner-group interleaving parameters according to another embodiment of the present invention.

FIG. 104 illustrates a write operation of inner group interleaving in NUC-1024 according to an embodiment of the present invention.

FIG. 105 illustrates a read operation of inner group interleaving in NUC-1024 according to an embodiment of the present invention.

FIG. 106 is a table illustrating inner-group interleaving parameters according to another embodiment of the present invention.

FIG. 107 illustrates a remaining QC block operation according to another embodiment of the present invention.

FIG. 108 is a block diagram illustrating a bit interleaver according to another embodiment of the present invention.

FIG. 109 corresponds to equations showing a process of obtaining a degree profile through density evolution according to an embodiment of the present invention.

FIG. 110 illustrates a degree profile according to an embodiment of the present invention.

FIG. 111 illustrates bit interleaving and bit deinterleaving according to another embodiment of the present invention.

FIG. 112 illustrates a bit interleaving pattern according to an embodiment of the present invention.

FIG. 113 illustrates a writing configuration according to an embodiment of the present invention.

FIG. 114 illustrates a cyclic shift operation according to an embodiment of the present invention.

FIG. 115 illustrates bit interleaving with a QC level according to an embodiment of the present invention.

FIG. 116 illustrates a cyclic shift operation with a QC level and a bit level according to an embodiment of the present invention.

FIG. 117 illustrates a cyclic shift operation for evenly dispersed parity parts according to an embodiment of the present invention.

FIG. 118 is a block diagram illustrating the demapping & decoding module 9020 according to another embodiment of the present invention.

FIG. 119 illustrates a DE result and a buffer associated with the DE result according to an embodiment of the present invention.

FIG. 120 is a graph illustrating bit error rate (BER) and block error rate (BLER) results of a symbol according to an embodiment of the present invention.

FIG. 121 is a flowchart illustrating a method for transmitting broadcast signals according to an embodiment of the present invention.

FIG. 122 is a flowchart illustrating a method for receiving broadcast signals according to an embodiment of the present invention.

DETAILED DESCRIPTION OF THE INVENTION

Reference will now be made in detail to the preferred embodiments of the present invention, examples of which are illustrated in the accompanying drawings. The detailed description, which will be given below with reference to the accompanying drawings, is intended to explain exemplary embodiments of the present invention, rather than to show the only embodiments that can be implemented according to the present invention. The following detailed description includes specific details in order to provide a thorough understanding of the present invention. However, it will be apparent to those skilled in the art that the present invention may be practiced without such specific details.

Although most terms used in the present invention have been selected from general ones widely used in the art, some terms have been arbitrarily selected by the applicant and their meanings are explained in detail in the following description as needed. Thus, the present invention should be understood based upon the intended meanings of the terms rather than their simple names or meanings.

The present invention provides apparatuses and methods for transmitting and receiving broadcast signals for future broadcast services. Future broadcast services according to an embodiment of the present invention include a terrestrial broadcast service, a mobile broadcast service, a UHDTV service, etc. The present invention may process broadcast signals for the future broadcast services through non-MIMO (Multiple Input Multiple Output) or MIMO according to one embodiment. A non-MIMO scheme according to an embodiment of the present invention may include a MISO (Multiple Input Single Output) scheme, a SISO (Single Input Single Output) scheme, etc.

While MISO or MIMO uses two antennas in the following for convenience of description, the present invention is applicable to systems using two or more antennas.

The present invention may defines three physical layer (PL) profiles—base, handheld and advanced profiles—each optimized to minimize receiver complexity while attaining the performance required for a particular use case. The physical layer (PHY) profiles are subsets of all configurations that a corresponding receiver should implement.

The three PHY profiles share most of the functional blocks but differ slightly in specific blocks and/or parameters. Additional PHY profiles can be defined in the future. For the system evolution, future profiles can also be multiplexed with the existing profiles in a single RF channel through a future extension frame (FEF). The details of each PHY profile are described below.

1. Base Profile

The base profile represents a main use case for fixed receiving devices that are usually connected to a roof-top antenna. The base profile also includes portable devices that could be transported to a place but belong to a relatively stationary reception category. Use of the base profile could be extended to handheld devices or even vehicular by some improved implementations, but those use cases are not expected for the base profile receiver operation.

Target SNR range of reception is from approximately 10 to 20 dB, which includes the 15 dB SNR reception capability of the existing broadcast system (e.g. ATSC A/53). The receiver complexity and power consumption is not as critical as in the battery-operated handheld devices, which will use the handheld profile. Key system parameters for the base profile are listed in below table 1.

TABLE 1 LDPC codeword length 16K, 64K bits Constellation size 4~10 bpcu (bits per channel use) Time de-interleaving memory size ≦2¹⁹ data cells Pilot patterns Pilot pattern for fixed reception FFT size 16K, 32K points

2. Handheld Profile

The handheld profile is designed for use in handheld and vehicular devices that operate with battery power. The devices can be moving with pedestrian or vehicle speed. The power consumption as well as the receiver complexity is very important for the implementation of the devices of the handheld profile. The target SNR range of the handheld profile is approximately 0 to 10 dB, but can be configured to reach below 0 dB when intended for deeper indoor reception.

In addition to low SNR capability, resilience to the Doppler Effect caused by receiver mobility is the most important performance attribute of the handheld profile. Key system parameters for the handheld profile are listed in the below table 2.

TABLE 2 LDPC codeword length 16K bits Constellation size 2~8 bpcu Time de-interleaving memory size ≦2¹⁸ data cells Pilot patterns Pilot patterns for mobile and indoor reception FFT size 8K, 16K points

3. Advanced profile

The advanced profile provides highest channel capacity at the cost of more implementation complexity. This profile requires using MIMO transmission and reception, and UHDTV service is a target use case for which this profile is specifically designed. The increased capacity can also be used to allow an increased number of services in a given bandwidth, e.g., multiple SDTV or HDTV services.

The target SNR range of the advanced profile is approximately 20 to 30 dB. MIMO transmission may initially use existing elliptically-polarized transmission equipment, with extension to full-power cross-polarized transmission in the future. Key system parameters for the advanced profile are listed in below table 3.

TABLE 3 LDPC codeword length 16K, 64K bits Constellation size 8~12 bpcu Time de-interleaving memory size ≦2¹⁹ data cells Pilot patterns Pilot pattern for fixed reception FFT size 16K, 32K points

In this case, the base profile can be used as a profile for both the terrestrial broadcast service and the mobile broadcast service. That is, the base profile can be used to define a concept of a profile which includes the mobile profile. Also, the advanced profile can be divided advanced profile for a base profile with MIMO and advanced profile for a handheld profile with MIMO. Moreover, the three profiles can be changed according to intention of the designer.

The following terms and definitions may apply to the present invention. The following terms and definitions can be changed according to design.

auxiliary stream: sequence of cells carrying data of as yet undefined modulation and coding, which may be used for future extensions or as required by broadcasters or network operators

base data pipe: data pipe that carries service signaling data

baseband frame (or BBFRAME): set of K_(bch) bits which form the input to one FEC encoding process (BCH and LDPC encoding)

cell: modulation value that is carried by one carrier of the OFDM transmission

coded block: LDPC-encoded block of PLS1 data or one of the LDPC-encoded blocks of PLS2 data

data pipe: logical channel in the physical layer that carries service data or related metadata, which may carry one or multiple service(s) or service component(s).

data pipe unit: a basic unit for allocating data cells to a DP in a frame.

data symbol: OFDM symbol in a frame which is not a preamble symbol (the frame signaling symbol and frame edge symbol is included in the data symbol)

DP_ID: this 8-bit field identifies uniquely a DP within the system identified by the SYSTEM_ID

dummy cell: cell carrying a pseudo-random value used to fill the remaining capacity not used for PLS signaling, DPs or auxiliary streams

emergency alert channel: part of a frame that carries EAS information data

frame: physical layer time slot that starts with a preamble and ends with a frame edge symbol

frame repetition unit: a set of frames belonging to same or different physical layer profile including a FEF, which is repeated eight times in a super-frame

fast information channel: a logical channel in a frame that carries the mapping information between a service and the corresponding base DP

FECBLOCK: set of LDPC-encoded bits of a DP data

FFT size: nominal FFT size used for a particular mode, equal to the active symbol period T_(S) expressed in cycles of the elementary period T

frame signaling symbol: OFDM symbol with higher pilot density used at the start of a frame in certain combinations of FFT size, guard interval and scattered pilot pattern, which carries a part of the PLS data

frame edge symbol: OFDM symbol with higher pilot density used at the end of a frame in certain combinations of FFT size, guard interval and scattered pilot pattern

frame-group: the set of all the frames having the same PHY profile type in a super-frame.

future extension frame: physical layer time slot within the super-frame that could be used for future extension, which starts with a preamble

Futurecast UTB system: proposed physical layer broadcasting system, of which the input is one or more MPEG2-TS or IP or general stream(s) and of which the output is an RF signal

input stream: A stream of data for an ensemble of services delivered to the end users by the system.

normal data symbol: data symbol excluding the frame signaling symbol and the frame edge symbol

PHY profile: subset of all configurations that a corresponding receiver should implement

PLS: physical layer signaling data consisting of PLS1 and PLS2

PLS1: a first set of PLS data carried in the FSS symbols having a fixed size, coding and modulation, which carries basic information about the system as well as the parameters needed to decode the PLS2

NOTE: PLS1 data remains constant for the duration of a frame-group.

PLS2: a second set of PLS data transmitted in the FSS symbol, which carries more detailed PLS data about the system and the DPs

PLS2 dynamic data: PLS2 data that may dynamically change frame-by-frame

PLS2 static data: PLS2 data that remains static for the duration of a frame-group

preamble signaling data: signaling data carried by the preamble symbol and used to identify the basic mode of the system

preamble symbol: fixed-length pilot symbol that carries basic PLS data and is located in the beginning of a frame

NOTE: The preamble symbol is mainly used for fast initial band scan to detect the system signal, its timing, frequency offset, and FFT-size.

reserved for future use: not defined by the present document but may be defined in future

super-frame: set of eight frame repetition units

time interleaving block (TI block): set of cells within which time interleaving is carried out, corresponding to one use of the time interleaver memory

TI group: unit over which dynamic capacity allocation for a particular DP is carried out, made up of an integer, dynamically varying number of XFECBLOCKs.

NOTE: The TI group may be mapped directly to one frame or may be mapped to multiple frames. It may contain one or more TI blocks.

Type 1 DP: DP of a frame where all DPs are mapped into the frame in TDM fashion

Type 2 DP: DP of a frame where all DPs are mapped into the frame in FDM fashion

XFECBLOCK: set of N_(cells) cells carrying all the bits of one LDPC FECBLOCK

FIG. 1 illustrates a structure of an apparatus for transmitting broadcast signals for future broadcast services according to an embodiment of the present invention.

The apparatus for transmitting broadcast signals for future broadcast services according to an embodiment of the present invention can include an input formatting block 1000, a BICM (Bit interleaved coding & modulation) block 1010, a frame structure block 1020, an OFDM (Orthogonal Frequency Division Multiplexing) generation block 1030 and a signaling generation block 1040. A description will be given of the operation of each module of the apparatus for transmitting broadcast signals.

IP stream/packets and MPEG2-TS are the main input formats, other stream types are handled as General Streams. In addition to these data inputs, Management Information is input to control the scheduling and allocation of the corresponding bandwidth for each input stream. One or multiple TS stream(s), IP stream(s) and/or General Stream(s) inputs are simultaneously allowed.

The input formatting block 1000 can demultiplex each input stream into one or multiple data pipe(s), to each of which an independent coding and modulation is applied. The data pipe (DP) is the basic unit for robustness control, thereby affecting quality-of-service (QoS). One or multiple service(s) or service component(s) can be carried by a single DP. Details of operations of the input formatting block 1000 will be described later.

The data pipe is a logical channel in the physical layer that carries service data or related metadata, which may carry one or multiple service(s) or service component(s).

Also, the data pipe unit: a basic unit for allocating data cells to a DP in a frame.

In the BICM block 1010, parity data is added for error correction and the encoded bit streams are mapped to complex-value constellation symbols. The symbols are interleaved across a specific interleaving depth that is used for the corresponding DP. For the advanced profile, MIMO encoding is performed in the BICM block 1010 and the additional data path is added at the output for MIMO transmission. Details of operations of the BICM block 1010 will be described later.

The Frame Building block 1020 can map the data cells of the input DPs into the OFDM symbols within a frame. After mapping, the frequency interleaving is used for frequency-domain diversity, especially to combat frequency-selective fading channels. Details of operations of the Frame Building block 1020 will be described later.

After inserting a preamble at the beginning of each frame, the OFDM Generation block 1030 can apply conventional OFDM modulation having a cyclic prefix as guard interval. For antenna space diversity, a distributed MISO scheme is applied across the transmitters. In addition, a Peak-to-Average Power Reduction (PAPR) scheme is performed in the time domain. For flexible network planning, this proposal provides a set of various FFT sizes, guard interval lengths and corresponding pilot patterns. Details of operations of the OFDM Generation block 1030 will be described later.

The Signaling Generation block 1040 can create physical layer signaling information used for the operation of each functional block. This signaling information is also transmitted so that the services of interest are properly recovered at the receiver side. Details of operations of the Signaling Generation block 1040 will be described later.

FIGS. 2, 3 and 4 illustrate the input formatting block 1000 according to embodiments of the present invention. A description will be given of each figure.

FIG. 2 illustrates an input formatting block according to one embodiment of the present invention. FIG. 2 shows an input formatting module when the input signal is a single input stream.

The input formatting block illustrated in FIG. 2 corresponds to an embodiment of the input formatting block 1000 described with reference to FIG. 1.

The input to the physical layer may be composed of one or multiple data streams. Each data stream is carried by one DP. The mode adaptation modules slice the incoming data stream into data fields of the baseband frame (BBF). The system supports three types of input data streams: MPEG2-TS, Internet protocol (IP) and Generic stream (GS). MPEG2-TS is characterized by fixed length (188 byte) packets with the first byte being a sync-byte (0x47). An IP stream is composed of variable length IP datagram packets, as signaled within IP packet headers. The system supports both IPv4 and IPv6 for the IP stream. GS may be composed of variable length packets or constant length packets, signaled within encapsulation packet headers.

(a) shows a mode adaptation block 2000 and a stream adaptation 2010 for signal DP and (b) shows a PLS generation block 2020 and a PLS scrambler 2030 for generating and processing PLS data. A description will be given of the operation of each block.

The Input Stream Splitter splits the input TS, IP, GS streams into multiple service or service component (audio, video, etc.) streams. The mode adaptation module 2010 is comprised of a CRC Encoder, BB (baseband) Frame Slicer, and BB Frame Header Insertion block.

The CRC Encoder provides three kinds of CRC encoding for error detection at the user packet (UP) level, i.e., CRC-8, CRC-16, and CRC-32. The computed CRC bytes are appended after the UP. CRC-8 is used for TS stream and CRC-32 for IP stream. If the GS stream doesn't provide the CRC encoding, the proposed CRC encoding should be applied.

BB Frame Slicer maps the input into an internal logical-bit format. The first received bit is defined to be the MSB. The BB Frame Slicer allocates a number of input bits equal to the available data field capacity. To allocate a number of input bits equal to the BBF payload, the UP packet stream is sliced to fit the data field of BBF.

BB Frame Header Insertion block can insert fixed length BBF header of 2 bytes is inserted in front of the BB Frame. The BBF header is composed of STUFFI (1 bit), SYNCD (13 bits), and RFU (2 bits). In addition to the fixed 2-Byte BBF header, BBF can have an extension field (1 or 3 bytes) at the end of the 2-byte BBF header.

The stream adaptation 2010 is comprised of stuffing insertion block and BB scrambler.

The stuffing insertion block can insert stuffing field into a payload of a BB frame. If the input data to the stream adaptation is sufficient to fill a BB-Frame, STUFFI is set to ‘0’ and the BBF has no stuffing field. Otherwise STUFFI is set to ‘1’ and the stuffing field is inserted immediately after the BBF header. The stuffing field comprises two bytes of the stuffing field header and a variable size of stuffing data.

The BB scrambler scrambles complete BBF for energy dispersal. The scrambling sequence is synchronous with the BBF. The scrambling sequence is generated by the feed-back shift register.

The PLS generation block 2020 can generate physical layer signaling (PLS) data. The PLS provides the receiver with a means to access physical layer DPs. The PLS data consists of PLS1 data and PLS2 data.

The PLS1 data is a first set of PLS data carried in the FSS symbols in the frame having a fixed size, coding and modulation, which carries basic information about the system as well as the parameters needed to decode the PLS2 data. The PLS1 data provides basic transmission parameters including parameters required to enable the reception and decoding of the PLS2 data. Also, the PLS1 data remains constant for the duration of a frame-group.

The PLS2 data is a second set of PLS data transmitted in the FSS symbol, which carries more detailed PLS data about the system and the DPs. The PLS2 contains parameters that provide sufficient information for the receiver to decode the desired DP. The PLS2 signaling further consists of two types of parameters, PLS2 Static data (PLS2-STAT data) and PLS2 dynamic data (PLS2-DYN data). The PLS2 Static data is PLS2 data that remains static for the duration of a frame-group and the PLS2 dynamic data is PLS2 data that may dynamically change frame-by-frame.

Details of the PLS data will be described later.

The PLS scrambler 2030 can scramble the generated PLS data for energy dispersal.

The above-described blocks may be omitted or replaced by blocks having similar or identical functions.

FIG. 3 illustrates an input formatting block according to another embodiment of the present invention.

The input formatting block illustrated in FIG. 3 corresponds to an embodiment of the input formatting block 1000 described with reference to FIG. 1.

FIG. 3 shows a mode adaptation block of the input formatting block when the input signal corresponds to multiple input streams.

The mode adaptation block of the input formatting block for processing the multiple input streams can independently process the multiple input streams.

Referring to FIG. 3, the mode adaptation block for respectively processing the multiple input streams can include an input stream splitter 3000, an input stream synchronizer 3010, a compensating delay block 3020, a null packet deletion block 3030, a head compression block 3040, a CRC encoder 3050, a BB frame slicer 3060 and a BB header insertion block 3070. Description will be given of each block of the mode adaptation block.

Operations of the CRC encoder 3050, BB frame slicer 3060 and BB header insertion block 3070 correspond to those of the CRC encoder, BB frame slicer and BB header insertion block described with reference to FIG. 2 and thus description thereof is omitted.

The input stream splitter 3000 can split the input TS, IP, GS streams into multiple service or service component (audio, video, etc.) streams.

The input stream synchronizer 3010 may be referred as ISSY. The ISSY can provide suitable means to guarantee Constant Bit Rate (CBR) and constant end-to-end transmission delay for any input data format. The ISSY is always used for the case of multiple DPs carrying TS, and optionally used for multiple DPs carrying GS streams.

The compensating delay block 3020 can delay the split TS packet stream following the insertion of ISSY information to allow a TS packet recombining mechanism without requiring additional memory in the receiver.

The null packet deletion block 3030, is used only for the TS input stream case. Some TS input streams or split TS streams may have a large number of null-packets present in order to accommodate VBR (variable bit-rate) services in a CBR TS stream. In this case, in order to avoid unnecessary transmission overhead, null-packets can be identified and not transmitted. In the receiver, removed null-packets can be re-inserted in the exact place where they were originally by reference to a deleted null-packet (DNP) counter that is inserted in the transmission, thus guaranteeing constant bit-rate and avoiding the need for time-stamp (PCR) updating.

The head compression block 3040 can provide packet header compression to increase transmission efficiency for TS or IP input streams. Because the receiver can have a priori information on certain parts of the header, this known information can be deleted in the transmitter.

For Transport Stream, the receiver has a-priori information about the sync-byte configuration (0x47) and the packet length (188 Byte). If the input TS stream carries content that has only one PID, i.e., for only one service component (video, audio, etc.) or service sub-component (SVC base layer, SVC enhancement layer, MVC base view or MVC dependent views), TS packet header compression can be applied (optionally) to the Transport Stream. IP packet header compression is used optionally if the input steam is an IP stream.

The above-described blocks may be omitted or replaced by blocks having similar or identical functions.

FIG. 4 illustrates an input formatting block according to another embodiment of the present invention.

The input formatting block illustrated in FIG. 4 corresponds to an embodiment of the input formatting block 1000 described with reference to FIG. 1.

FIG. 4 illustrates a stream adaptation block of the input formatting module when the input signal corresponds to multiple input streams.

Referring to FIG. 4, the mode adaptation block for respectively processing the multiple input streams can include a scheduler 4000, an 1-Frame delay block 4010, a stuffing insertion block 4020, an in-band signaling 4030, a BB Frame scrambler 4040, a PLS generation block 4050 and a PLS scrambler 4060. Description will be given of each block of the stream adaptation block.

Operations of the stuffing insertion block 4020, the BB Frame scrambler 4040, the PLS generation block 4050 and the PLS scrambler 4060 correspond to those of the stuffing insertion block, BB scrambler, PLS generation block and the PLS scrambler described with reference to FIG. 2 and thus description thereof is omitted.

The scheduler 4000 can determine the overall cell allocation across the entire frame from the amount of FECBLOCKs of each DP. Including the allocation for PLS, EAC and FIC, the scheduler generate the values of PLS2-DYN data, which is transmitted as in-band signaling or PLS cell in FSS of the frame. Details of FECBLOCK, EAC and FIC will be described later.

The 1-Frame delay block 4010 can delay the input data by one transmission frame such that scheduling information about the next frame can be transmitted through the current frame for in-band signaling information to be inserted into the DPs.

The in-band signaling 4030 can insert un-delayed part of the PLS2 data into a DP of a frame.

The above-described blocks may be omitted or replaced by blocks having similar or identical functions.

FIG. 5 illustrates a BICM block according to an embodiment of the present invention.

The BICM block illustrated in FIG. 5 corresponds to an embodiment of the BICM block 1010 described with reference to FIG. 1.

As described above, the apparatus for transmitting broadcast signals for future broadcast services according to an embodiment of the present invention can provide a terrestrial broadcast service, mobile broadcast service, UHDTV service, etc.

Since QoS (quality of service) depends on characteristics of a service provided by the apparatus for transmitting broadcast signals for future broadcast services according to an embodiment of the present invention, data corresponding to respective services needs to be processed through different schemes. Accordingly, the a BICM block according to an embodiment of the present invention can independently process DPs input thereto by independently applying SISO, MISO and MIMO schemes to the data pipes respectively corresponding to data paths. Consequently, the apparatus for transmitting broadcast signals for future broadcast services according to an embodiment of the present invention can control QoS for each service or service component transmitted through each DP.

(a) shows the BICM block shared by the base profile and the handheld profile and (b) shows the BICM block of the advanced profile.

The BICM block shared by the base profile and the handheld profile and the BICM block of the advanced profile can include plural processing blocks for processing each DP.

A description will be given of each processing block of the BICM block for the base profile and the handheld profile and the BICM block for the advanced profile.

A processing block 5000 of the BICM block for the base profile and the handheld profile can include a Data FEC encoder 5010, a bit interleaver 5020, a constellation mapper 5030, an SSD (Signal Space Diversity) encoding block 5040 and a time interleaver 5050.

The Data FEC encoder 5010 can perform the FEC encoding on the input BBF to generate FECBLOCK procedure using outer coding (BCH), and inner coding (LDPC). The outer coding (BCH) is optional coding method. Details of operations of the Data FEC encoder 5010 will be described later.

The bit interleaver 5020 can interleave outputs of the Data FEC encoder 5010 to achieve optimized performance with combination of the LDPC codes and modulation scheme while providing an efficiently implementable structure. Details of operations of the bit interleaver 5020 will be described later.

The constellation mapper 5030 can modulate each cell word from the bit interleaver 5020 in the base and the handheld profiles, or cell word from the Cell-word demultiplexer 5010-1 in the advanced profile using either QPSK, QAM-16, non-uniform QAM (NUQ-64, NUQ-256, NUQ-1024) or non-uniform constellation (NUC-16, NUC-64, NUC-256, NUC-1024) to give a power-normalized constellation point, e_(l). This constellation mapping is applied only for DPs. Observe that QAM-16 and NUQs are square shaped, while NUCs have arbitrary shape. When each constellation is rotated by any multiple of 90 degrees, the rotated constellation exactly overlaps with its original one. This “rotation-sense” symmetric property makes the capacities and the average powers of the real and imaginary components equal to each other. Both NUQs and NUCs are defined specifically for each code rate and the particular one used is signaled by the parameter DP_MOD filed in PLS2 data.

The SSD encoding block 5040 can precode cells in two (2D), three (3D), and four (4D) dimensions to increase the reception robustness under difficult fading conditions.

The time interleaver 5050 can operates at the DP level. The parameters of time interleaving (TI) may be set differently for each DP. Details of operations of the time interleaver 5050 will be described later.

A processing block 5000-1 of the BICM block for the advanced profile can include the Data FEC encoder, bit interleaver, constellation mapper, and time interleaver. However, the processing block 5000-1 is distinguished from the processing block 5000 further includes a cell-word demultiplexer 5010-1 and a MIMO encoding block 5020-1.

Also, the operations of the Data FEC encoder, bit interleaver, constellation mapper, and time interleaver in the processing block 5000-1 correspond to those of the Data FEC encoder 5010, bit interleaver 5020, constellation mapper 5030, and time interleaver 5050 described and thus description thereof is omitted.

The cell-word demultiplexer 5010-1 is used for the DP of the advanced profile to divide the single cell-word stream into dual cell-word streams for MIMO processing. Details of operations of the cell-word demultiplexer 5010-1 will be described later.

The MIMO encoding block 5020-1 can processing the output of the cell-word demultiplexer 5010-1 using MIMO encoding scheme. The MIMO encoding scheme was optimized for broadcasting signal transmission. The MIMO technology is a promising way to get a capacity increase but it depends on channel characteristics. Especially for broadcasting, the strong LOS component of the channel or a difference in the received signal power between two antennas caused by different signal propagation characteristics makes it difficult to get capacity gain from MIMO. The proposed MIMO encoding scheme overcomes this problem using a rotation-based pre-coding and phase randomization of one of the MIMO output signals.

MIMO encoding is intended for a 2×2 MIMO system requiring at least two antennas at both the transmitter and the receiver. Two MIMO encoding modes are defined in this proposal; full-rate spatial multiplexing (FR-SM) and full-rate full-diversity spatial multiplexing (FRFD-SM). The FR-SM encoding provides capacity increase with relatively small complexity increase at the receiver side while the FRFD-SM encoding provides capacity increase and additional diversity gain with a great complexity increase at the receiver side. The proposed MIMO encoding scheme has no restriction on the antenna polarity configuration.

MIMO processing is required for the advanced profile frame, which means all DPs in the advanced profile frame are processed by the MIMO encoder. MIMO processing is applied at DP level. Pairs of the Constellation Mapper outputs NUQ (e_(1,i) and e_(2,i)) are fed to the input of the MIMO Encoder. Paired MIMO Encoder output (g1,i and g2,i) is transmitted by the same carrier k and OFDM symbol l of their respective TX antennas.

The above-described blocks may be omitted or replaced by blocks having similar or identical functions.

FIG. 6 illustrates a BICM block according to another embodiment of the present invention.

The BICM block illustrated in FIG. 6 corresponds to an embodiment of the BICM block 1010 described with reference to FIG. 1.

FIG. 6 illustrates a BICM block for protection of physical layer signaling (PLS), emergency alert channel (EAC) and fast information channel (FIC). EAC is a part of a frame that carries EAS information data and FIC is a logical channel in a frame that carries the mapping information between a service and the corresponding base DP. Details of the EAC and FIC will be described later.

Referring to FIG. 6, the BICM block for protection of PLS, EAC and FIC can include a PLS FEC encoder 6000, a bit interleaver 6010, a constellation mapper 6020 and time interleaver 6030.

Also, the PLS FEC encoder 6000 can include a scrambler, BCH encoding/zero insertion block, LDPC encoding block and LDPC parity punturing block. Description will be given of each block of the BICM block.

The PLS FEC encoder 6000 can encode the scrambled PLS 1/2 data, EAC and FIC section.

The scrambler can scramble PLS1 data and PLS2 data before BCH encoding and shortened and punctured LDPC encoding.

The BCH encoding/zero insertion block can perform outer encoding on the scrambled PLS 1/2 data using the shortened BCH code for PLS protection and insert zero bits after the BCH encoding. For PLS1 data only, the output bits of the zero insertion may be permutted before LDPC encoding.

The LDPC encoding block can encode the output of the BCH encoding/zero insertion block using LDPC code. To generate a complete coded block, C_(ldpc), parity bits, P_(ldpc) are encoded systematically from each zero-inserted PLS information block, I_(ldpc) and appended after it.

C _(ldpc) =[I _(ldpc) P _(ldpc) ]=[i ₀ ,i ₁ , . . . ,i _(K) _(ldpc) ⁻¹ ,p ₀ ,p ₁ , . . . ,p _(N) _(ldpc) _(−K) _(ldpc) ⁻¹]  [Expression 1]

The LDPC code parameters for PLS1 and PLS2 are as following table 4.

TABLE 4 Signal- ling N_(bch)_ K_(ldpc) code Type K_(sig) K_(bch) _(parity) (=N_(bch)) N_(ldpc) N_(ldpc)_parity rate Q_(ldpc) PLS1 342 1020 60 1080 4320 3240 ¼ 36 PLS2 <1021 2100 2160 7200 5040 3/10 56 >1020

The LDPC parity punturing block can perform puncturing on the PLS1 data and PLS 2 data.

When shortening is applied to the PLS1 data protection, some LDPC parity bits are punctured after LDPC encoding. Also, for the PLS2 data protection, the LDPC parity bits of PLS2 are punctured after LDPC encoding. These punctured bits are not transmitted.

The bit interleaver 6010 can interleave the each shortened and punctured PLS1 data and PLS2 data.

The constellation mapper 6020 can map the bit interleaved PLS1 data and PLS2 data onto constellations.

The time interleaver 6030 can interleave the mapped PLS1 data and PLS2 data.

The above-described blocks may be omitted or replaced by blocks having similar or identical functions.

FIG. 7 illustrates a frame building block according to one embodiment of the present invention.

The frame building block illustrated in FIG. 7 corresponds to an embodiment of the frame building block 1020 described with reference to FIG. 1.

Referring to FIG. 7, the frame building block can include a delay compensation block 7000, a cell mapper 7010 and a frequency interleaver 7020. Description will be given of each block of the frame building block.

The delay compensation block 7000 can adjust the timing between the data pipes and the corresponding PLS data to ensure that they are co-timed at the transmitter end. The PLS data is delayed by the same amount as data pipes are by addressing the delays of data pipes caused by the Input Formatting block and BICM block. The delay of the BICM block is mainly due to the time interleaver. In-band signaling data carries information of the next TI group so that they are carried one frame ahead of the DPs to be signaled. The Delay Compensating block delays in-band signaling data accordingly.

The cell mapper 7010 can map PLS, EAC, FIC, DPs, auxiliary streams and dummy cells into the active carriers of the OFDM symbols in the frame. The basic function of the cell mapper 7010 is to map data cells produced by the TIs for each of the DPs, PLS cells, and EAC/FIC cells, if any, into arrays of active OFDM cells corresponding to each of the OFDM symbols within a frame. Service signaling data (such as PSI (program specific information)/SI) can be separately gathered and sent by a data pipe. The Cell Mapper operates according to the dynamic information produced by the scheduler and the configuration of the frame structure. Details of the frame will be described later.

The frequency interleaver 7020 can randomly interleave data cells received from the cell mapper 7010 to provide frequency diversity. Also, the frequency interleaver 7020 can operate on very OFDM symbol pair comprised of two sequential OFDM symbols using a different interleaving-seed order to get maximum interleaving gain in a single frame.

The above-described blocks may be omitted or replaced by blocks having similar or identical functions.

FIG. 8 illustrates an OFDM generation block according to an embodiment of the present invention.

The OFDM generation block illustrated in FIG. 8 corresponds to an embodiment of the OFDM generation block 1030 described with reference to FIG. 1.

The OFDM generation block modulates the OFDM carriers by the cells produced by the Frame Building block, inserts the pilots, and produces the time domain signal for transmission. Also, this block subsequently inserts guard intervals, and applies PAPR (Peak-to-Average Power Radio) reduction processing to produce the final RF signal.

Referring to FIG. 8, the OFDM generation block can include a pilot and reserved tone insertion block 8000, a 2D-eSFN encoding block 8010, an IFFT (Inverse Fast Fourier Transform) block 8020, a PAPR reduction block 8030, a guard interval insertion block 8040, a preamble insertion block 8050, other system insertion block 8060 and a DAC block 8070. Description will be given of each block of the frame building block.

The pilot and reserved tone insertion block 8000 can insert pilots and the reserved tone.

Various cells within the OFDM symbol are modulated with reference information, known as pilots, which have transmitted values known a priori in the receiver. The information of pilot cells is made up of scattered pilots, continual pilots, edge pilots, FSS (frame signaling symbol) pilots and FES (frame edge symbol) pilots. Each pilot is transmitted at a particular boosted power level according to pilot type and pilot pattern. The value of the pilot information is derived from a reference sequence, which is a series of values, one for each transmitted carrier on any given symbol. The pilots can be used for frame synchronization, frequency synchronization, time synchronization, channel estimation, and transmission mode identification, and also can be used to follow the phase noise.

Reference information, taken from the reference sequence, is transmitted in scattered pilot cells in every symbol except the preamble, FSS and FES of the frame. Continual pilots are inserted in every symbol of the frame. The number and location of continual pilots depends on both the FFT size and the scattered pilot pattern. The edge carriers are edge pilots in every symbol except for the preamble symbol. They are inserted in order to allow frequency interpolation up to the edge of the spectrum. FSS pilots are inserted in FSS(s) and FES pilots are inserted in FES. They are inserted in order to allow time interpolation up to the edge of the frame.

The system according to an embodiment of the present invention supports the SFN network, where distributed MISO scheme is optionally used to support very robust transmission mode. The 2D-eSFN is a distributed MISO scheme that uses multiple TX antennas, each of which is located in the different transmitter site in the SFN network.

The 2D-eSFN encoding block 8010 can process a 2D-eSFN processing to distorts the phase of the signals transmitted from multiple transmitters, in order to create both time and frequency diversity in the SFN configuration. Hence, burst errors due to low flat fading or deep-fading for a long time can be mitigated.

The IFFT block 8020 can modulate the output from the 2D-eSFN encoding block 8010 using OFDM modulation scheme. Any cell in the data symbols which has not been designated as a pilot (or as a reserved tone) carries one of the data cells from the frequency interleaver. The cells are mapped to OFDM carriers.

The PAPR reduction block 8030 can perform a PAPR reduction on input signal using various PAPR reduction algorithm in the time domain.

The guard interval insertion block 8040 can insert guard intervals and the preamble insertion block 8050 can insert preamble in front of the signal. Details of a structure of the preamble will be described later. The other system insertion block 8060 can multiplex signals of a plurality of broadcast transmission/reception systems in the time domain such that data of two or more different broadcast transmission/reception systems providing broadcast services can be simultaneously transmitted in the same RF signal bandwidth. In this case, the two or more different broadcast transmission/reception systems refer to systems providing different broadcast services. The different broadcast services may refer to a terrestrial broadcast service, mobile broadcast service, etc. Data related to respective broadcast services can be transmitted through different frames.

The DAC block 8070 can convert an input digital signal into an analog signal and output the analog signal. The signal output from the DAC block 7800 can be transmitted through multiple output antennas according to the physical layer profiles. A Tx antenna according to an embodiment of the present invention can have vertical or horizontal polarity.

The above-described blocks may be omitted or replaced by blocks having similar or identical functions according to design.

FIG. 9 illustrates a structure of an apparatus for receiving broadcast signals for future broadcast services according to an embodiment of the present invention.

The apparatus for receiving broadcast signals for future broadcast services according to an embodiment of the present invention can correspond to the apparatus for transmitting broadcast signals for future broadcast services, described with reference to FIG. 1.

The apparatus for receiving broadcast signals for future broadcast services according to an embodiment of the present invention can include a synchronization & demodulation module 9000, a frame parsing module 9010, a demapping & decoding module 9020, an output processor 9030 and a signaling decoding module 9040. A description will be given of operation of each module of the apparatus for receiving broadcast signals.

The synchronization & demodulation module 9000 can receive input signals through m Rx antennas, perform signal detection and synchronization with respect to a system corresponding to the apparatus for receiving broadcast signals and carry out demodulation corresponding to a reverse procedure of the procedure performed by the apparatus for transmitting broadcast signals.

The frame parsing module 9010 can parse input signal frames and extract data through which a service selected by a user is transmitted. If the apparatus for transmitting broadcast signals performs interleaving, the frame parsing module 9010 can carry out deinterleaving corresponding to a reverse procedure of interleaving. In this case, the positions of a signal and data that need to be extracted can be obtained by decoding data output from the signaling decoding module 9040 to restore scheduling information generated by the apparatus for transmitting broadcast signals.

The demapping & decoding module 9020 can convert the input signals into bit domain data and then deinterleave the same as necessary. The demapping & decoding module 9020 can perform demapping for mapping applied for transmission efficiency and correct an error generated on a transmission channel through decoding. In this case, the demapping & decoding module 9020 can obtain transmission parameters necessary for demapping and decoding by decoding the data output from the signaling decoding module 9040.

The output processor 9030 can perform reverse procedures of various compression/signal processing procedures which are applied by the apparatus for transmitting broadcast signals to improve transmission efficiency. In this case, the output processor 9030 can acquire necessary control information from data output from the signaling decoding module 9040. The output of the output processor 8300 corresponds to a signal input to the apparatus for transmitting broadcast signals and may be MPEG-TSs, IP streams (v4 or v6) and generic streams.

The signaling decoding module 9040 can obtain PLS information from the signal demodulated by the synchronization & demodulation module 9000. As described above, the frame parsing module 9010, demapping & decoding module 9020 and output processor 9030 can execute functions thereof using the data output from the signaling decoding module 9040.

FIG. 10 illustrates a frame structure according to an embodiment of the present invention.

FIG. 10 shows an example configuration of the frame types and FRUs in a super-frame. (a) shows a super frame according to an embodiment of the present invention, (b) shows FRU (Frame Repetition Unit) according to an embodiment of the present invention, (c) shows frames of variable PHY profiles in the FRU and (d) shows a structure of a frame.

A super-frame may be composed of eight FRUs. The FRU is a basic multiplexing unit for TDM of the frames, and is repeated eight times in a super-frame.

Each frame in the FRU belongs to one of the PHY profiles, (base, handheld, advanced) or FEE. The maximum allowed number of the frames in the FRU is four and a given PHY profile can appear any number of times from zero times to four times in the FRU (e.g., base, base, handheld, advanced). PHY profile definitions can be extended using reserved values of the PHY_PROFILE in the preamble, if required.

The FEF part is inserted at the end of the FRU, if included. When the FEF is included in the FRU, the minimum number of FEFs is 8 in a super-frame. It is not recommended that FEF parts be adjacent to each other.

One frame is further divided into a number of OFDM symbols and a preamble. As shown in (d), the frame comprises a preamble, one or more frame signaling symbols (FSS), normal data symbols and a frame edge symbol (FES).

The preamble is a special symbol that enables fast Futurecast UTB system signal detection and provides a set of basic transmission parameters for efficient transmission and reception of the signal. The detailed description of the preamble will be will be described later.

The main purpose of the FSS(s) is to carry the PLS data. For fast synchronization and channel estimation, and hence fast decoding of PLS data, the FSS has more dense pilot pattern than the normal data symbol. The FES has exactly the same pilots as the FSS, which enables frequency-only interpolation within the FES and temporal interpolation, without extrapolation, for symbols immediately preceding the FES.

FIG. 11 illustrates a signaling hierarchy structure of the frame according to an embodiment of the present invention.

FIG. 11 illustrates the signaling hierarchy structure, which is split into three main parts: the preamble signaling data 11000, the PLS1 data 11010 and the PLS2 data 11020. The purpose of the preamble, which is carried by the preamble symbol in every frame, is to indicate the transmission type and basic transmission parameters of that frame. The PLS1 enables the receiver to access and decode the PLS2 data, which contains the parameters to access the DP of interest. The PLS2 is carried in every frame and split into two main parts: PLS2-STAT data and PLS2-DYN data. The static and dynamic portion of PLS2 data is followed by padding, if necessary.

FIG. 12 illustrates preamble signaling data according to an embodiment of the present invention.

Preamble signaling data carries 21 bits of information that are needed to enable the receiver to access PLS data and trace DPs within the frame structure. Details of the preamble signaling data are as follows:

PHY_PROFILE: This 3-bit field indicates the PHY profile type of the current frame. The mapping of different PHY profile types is given in below table 5.

TABLE 5 Value PHY profile 000 Base profile 001 Handheld profile 010 Advanced profiled 011~110 Reserved 111 FEF

FFT_SIZE: This 2 bit field indicates the FFT size of the current frame within a frame-group, as described in below table 6.

TABLE 6 Value FFT size 00  8K FFT 01 16K FFT 10 32K FFT 11 Reserved

GI_FRACTION: This 3 bit field indicates the guard interval fraction value in the current super-frame, as described in below table 7.

TABLE 7 Value GI_FRACTION 000 ⅕ 001 1/10 010 1/20 011 1/40 100 1/80 101 1/160 110~111 Reserved

EAC_FLAG: This 1 bit field indicates whether the EAC is provided in the current frame. If this field is set to ‘1’, emergency alert service (EAS) is provided in the current frame. If this field set to ‘0’, EAS is not carried in the current frame. This field can be switched dynamically within a super-frame.

PILOT_MODE: This 1-bit field indicates whether the pilot mode is mobile mode or fixed mode for the current frame in the current frame-group. If this field is set to ‘0’, mobile pilot mode is used. If the field is set to ‘1’, the fixed pilot mode is used.

PAPR_FLAG: This 1-bit field indicates whether PAPR reduction is used for the current frame in the current frame-group. If this field is set to value ‘1’, tone reservation is used for PAPR reduction. If this field is set to ‘0’, PAPR reduction is not used.

FRU_CONFIGURE: This 3-bit field indicates the PHY profile type configurations of the frame repetition units (FRU) that are present in the current super-frame. All profile types conveyed in the current super-frame are identified in this field in all preambles in the current super-frame. The 3-bit field has a different definition for each profile, as show in below table 8.

TABLE 8 Current Current Current PHY_ PHY_ Current PHY_ PROFILE = PROFILE = PHY_ PROFILE = ′001′ ′010′ PROFILE = ′000′ (base) (handheld) (advanced) ′111′ (FEF) FRU_CONFIGURE = Only base Only handheld Only advanced Only FEF 000 profile profile present profile present present present FRU_CONFIGURE = Handheld profile Base profile Base profile Base profile 1XX present present present present FRU_CONFIGURE = Advanced Advanced Handheld profile Handheld profile X1X profile profile present present present present FRU_CONFIGURE = FEF FEF FEF Advanced XX1 present present present profile present

RESERVED: This 7-bit field is reserved for future use.

FIG. 13 illustrates PLS1 data according to an embodiment of the present invention.

PLS1 data provides basic transmission parameters including parameters required to enable the reception and decoding of the PLS2. As above mentioned, the PLS1 data remain unchanged for the entire duration of one frame-group. The detailed definition of the signaling fields of the PLS1 data are as follows:

PREAMBLE_DATA: This 20-bit field is a copy of the preamble signaling data excluding the EAC_FLAG.

NUM_FRAME_FRU: This 2-bit field indicates the number of the frames per FRU.

PAYLOAD_TYPE: This 3-bit field indicates the format of the payload data carried in the frame-group. PAYLOAD_TYPE is signaled as shown in table 9.

TABLE 9 value Payload type 1XX TS stream is transmitted X1X IP stream is transmitted XX1 GS stream is transmitted

NUM_FSS: This 2-bit field indicates the number of FSS symbols in the current frame.

SYSTEM_VERSION: This 8-bit field indicates the version of the transmitted signal format. The SYSTEM_VERSION is divided into two 4-bit fields, which are a major version and a minor version.

Major version: The MSB four bits of SYSTEM_VERSION field indicate major version information. A change in the major version field indicates a non-backward-compatible change. The default value is ‘0000’. For the version described in this standard, the value is set to ‘0000’.

Minor version: The LSB four bits of SYSTEM_VERSION field indicate minor version information. A change in the minor version field is backward-compatible.

CELL_ID: This is a 16-bit field which uniquely identifies a geographic cell in an ATSC network. An ATSC cell coverage area may consist of one or more frequencies, depending on the number of frequencies used per Futurecast UTB system. If the value of the CELL_ID is not known or unspecified, this field is set to ‘0’.

NETWORK_ID: This is a 16-bit field which uniquely identifies the current ATSC network.

SYSTEM_ID: This 16-bit field uniquely identifies the Futurecast UTB system within the ATSC network. The Futurecast UTB system is the terrestrial broadcast system whose input is one or more input streams (TS, IP, GS) and whose output is an RF signal. The Futurecast UTB system carries one or more PHY profiles and FEF, if any. The same Futurecast UTB system may carry different input streams and use different RF frequencies in different geographical areas, allowing local service insertion. The frame structure and scheduling is controlled in one place and is identical for all transmissions within a Futurecast UTB system. One or more Futurecast UTB systems may have the same SYSTEM_ID meaning that they all have the same physical layer structure and configuration.

The following loop consists of FRU_PHY_PROFILE, FRU_FRAME_LENGTH, FRU_GI_FRACTION, and RESERVED which are used to indicate the FRU configuration and the length of each frame type. The loop size is fixed so that four PHY profiles (including a FEF) are signaled within the FRU. If NUM_FRAME_FRU is less than 4, the unused fields are filled with zeros.

FRU_PHY_PROFILE: This 3-bit field indicates the PHY profile type of the (i+1)^(th) (i is the loop index) frame of the associated FRU. This field uses the same signaling format as shown in the table 8.

FRU_FRAME_LENGTH: This 2-bit field indicates the length of the (i+1)^(th) frame of the associated FRU. Using FRU_FRAME_LENGTH together with FRU_GI_FRACTION, the exact value of the frame duration can be obtained.

FRU_GI_FRACTION: This 3-bit field indicates the guard interval fraction value of the (i+1)^(th) frame of the associated FRU. FRU_GI_FRACTION is signaled according to the table 7.

RESERVED: This 4-bit field is reserved for future use.

The following fields provide parameters for decoding the PLS2 data.

PLS2_FEC_TYPE: This 2-bit field indicates the FEC type used by the PLS2 protection. The FEC type is signaled according to table 10. The details of the LDPC codes will be described later.

TABLE 10 Content PLS2 FEC type 00 4K-1/4 and 7K-3/10 LDPC codes 01~11 Reserved

PLS2_MOD: This 3-bit field indicates the modulation type used by the PLS2. The modulation type is signaled according to table 11.

TABLE 11 Value PLS2_MODE 000 BPSK 001 QPSK 010 QAM-16 011 NUQ-64 100~111 Reserved

PLS2_SIZE_CELL: This 15-bit field indicates C_(total) _(_) _(partial) _(_) _(block), the size (specified as the number of QAM cells) of the collection of full coded blocks for PLS2 that is carried in the current frame-group. This value is constant during the entire duration of the current frame-group.

PLS2_STAT_SIZE_BIT: This 14-bit field indicates the size, in bits, of the PLS2-STAT for the current frame-group. This value is constant during the entire duration of the current frame-group.

PLS2_DYN_SIZE_BIT: This 14-bit field indicates the size, in bits, of the PLS2-DYN for the current frame-group. This value is constant during the entire duration of the current frame-group.

PLS2_REP_FLAG: This 1-bit flag indicates whether the PLS2 repetition mode is used in the current frame-group. When this field is set to value ‘1’, the PLS2 repetition mode is activated. When this field is set to value ‘0’, the PLS2 repetition mode is deactivated.

PLS2_REP_SIZE_CELL: This 15-bit field indicates C_(total) _(_) _(partial) _(_) _(block), the size (specified as the number of QAM cells) of the collection of partial coded blocks for PLS2 carried in every frame of the current frame-group, when PLS2 repetition is used. If repetition is not used, the value of this field is equal to 0. This value is constant during the entire duration of the current frame-group.

PLS2_NEXT_FEC_TYPE: This 2-bit field indicates the FEC type used for PLS2 that is carried in every frame of the next frame-group. The FEC type is signaled according to the table 10.

PLS2_NEXT_MOD: This 3-bit field indicates the modulation type used for PLS2 that is carried in every frame of the next frame-group. The modulation type is signaled according to the table 11.

PLS 2_NEXT_REP_FLAG: This 1-bit flag indicates whether the PLS2 repetition mode is used in the next frame-group. When this field is set to value ‘1’, the PLS2 repetition mode is activated. When this field is set to value ‘0’, the PLS2 repetition mode is deactivated.

PLS2_NEXT_REP_SIZE_CELL: This 15-bit field indicates C_(total) _(_) _(full) _(_) _(block), The size (specified as the number of QAM cells) of the collection of full coded blocks for PLS2 that is carried in every frame of the next frame-group, when PLS2 repetition is used. If repetition is not used in the next frame-group, the value of this field is equal to 0. This value is constant during the entire duration of the current frame-group.

PLS2_NEXT_REP_STAT_SIZE_BIT: This 14-bit field indicates the size, in bits, of the PLS2-STAT for the next frame-group. This value is constant in the current frame-group.

PLS2_NEXT_REP_DYN_SIZE_BIT: This 14-bit field indicates the size, in bits, of the PLS2-DYN for the next frame-group. This value is constant in the current frame-group.

PLS2_AP_MODE: This 2-bit field indicates whether additional parity is provided for PLS2 in the current frame-group. This value is constant during the entire duration of the current frame-group. The below table 12 gives the values of this field. When this field is set to ‘00’, additional parity is not used for the PLS2 in the current frame-group.

TABLE 12 Value PLS2-AP mode 00 AP is not provided 01 AP1 mode 10~11 Reserved

PLS2_AP_SIZE_CELL: This 15-bit field indicates the size (specified as the number of QAM cells) of the additional parity bits of the PLS2. This value is constant during the entire duration of the current frame-group.

PLS2_NEXT_AP_MODE: This 2-bit field indicates whether additional parity is provided for PLS2 signaling in every frame of next frame-group. This value is constant during the entire duration of the current frame-group. The table 12 defines the values of this field

PLS2_NEXT_AP_SIZE_CELL: This 15-bit field indicates the size (specified as the number of QAM cells) of the additional parity bits of the PLS2 in every frame of the next frame-group. This value is constant during the entire duration of the current frame-group.

RESERVED: This 32-bit field is reserved for future use.

CRC_32: A 32-bit error detection code, which is applied to the entire PLS1 signaling.

FIG. 14 illustrates PLS2 data according to an embodiment of the present invention.

FIG. 14 illustrates PLS2-STAT data of the PLS2 data. The PLS2-STAT data are the same within a frame-group, while the PLS2-DYN data provide information that is specific for the current frame.

The details of fields of the PLS2-STAT data are as follows:

FIC_FLAG: This 1-bit field indicates whether the FIC is used in the current frame-group. If this field is set to ‘1’, the FIC is provided in the current frame. If this field set to ‘0’, the FIC is not carried in the current frame. This value is constant during the entire duration of the current frame-group.

AUX_FLAG: This 1-bit field indicates whether the auxiliary stream(s) is used in the current frame-group. If this field is set to ‘1’, the auxiliary stream is provided in the current frame. If this field set to ‘0’, the auxiliary stream is not carried in the current frame. This value is constant during the entire duration of current frame-group.

NUM_DP: This 6-bit field indicates the number of DPs carried within the current frame. The value of this field ranges from 1 to 64, and the number of DPs is NUM_DP+1.

DP_ID: This 6-bit field identifies uniquely a DP within a PHY profile.

DP_TYPE: This 3-bit field indicates the type of the DP. This is signaled according to the below table 13.

TABLE 13 Value DP Type 000 DP Type 1 001 DP Type 2 010~111 reserved

DP_GROUP_ID: This 8-bit field identifies the DP group with which the current DP is associated. This can be used by a receiver to access the DPs of the service components associated with a particular service, which will have the same DP_GROUP_ID.

BASE_DP_ID: This 6-bit field indicates the DP carrying service signaling data (such as PSI/SI) used in the Management layer. The DP indicated by BASE_DP_ID may be either a normal DP carrying the service signaling data along with the service data or a dedicated DP carrying only the service signaling data

DP_FEC_TYPE: This 2-bit field indicates the FEC type used by the associated DP. The FEC type is signaled according to the below table 14.

TABLE 14 Value FEC_TYPE 00 16K LDPC 01 64K LDPL 10~11 Reserved

DP_COD: This 4-bit field indicates the code rate used by the associated DP. The code rate is signaled according to the below table 15.

TABLE 15 Value Code rate 0000  5/15 0001  6/15 0010  7/15 0011  8/15 0100  9/15 0101 10/15 0110 11/15 0111 12/15 1000 13/15 1001~1111 Reserved

DP_MOD: This 4-bit field indicates the modulation used by the associated DP. The modulation is signaled according to the below table 16.

TABLE 16 Value Modulation 0000 QPSK 0001 QAM-16 0010 NUQ-64 0011 NUQ-256 0100 NUQ-1024 0101 NUC-16 0110 NUC-64 0111 NUC-256 1000 NUC-1024 1001~1111 reserved

DP_SSD_FLAG: This 1-bit field indicates whether the SSD mode is used in the associated DP. If this field is set to value ‘1’, SSD is used. If this field is set to value ‘0’, SSD is not used.

The following field appears only if PHY_PROFILE is equal to ‘010’, which indicates the advanced profile:

DP_MIMO: This 3-bit field indicates which type of MIMO encoding process is applied to the associated DP. The type of MIMO encoding process is signaled according to the table 17.

TABLE 17 Value MIMO encoding 000 FR-SM 001 FRFD-SM 010~111 reserved

DP_TI_TYPE: This 1-bit field indicates the type of time-interleaving. A value of ‘0’ indicates that one TI group corresponds to one frame and contains one or more TI-blocks. A value of ‘1’ indicates that one TI group is carried in more than one frame and contains only one TI-block.

DP_TI_LENGTH: The use of this 2-bit field (the allowed values are only 1, 2, 4, 8) is determined by the values set within the DP_TI_TYPE field as follows:

If the DP_TI_TYPE is set to the value ‘1’, this field indicates P_(I), the number of the frames to which each TI group is mapped, and there is one TI-block per TI group (N_(TI)=1). The allowed P_(I) values with 2-bit field are defined in the below table 18.

If the DP_TI_TYPE is set to the value ‘0’, this field indicates the number of TI-blocks N_(TI) per TI group, and there is one TI group per frame (P_(I)=1). The allowed P_(I) values with 2-bit field are defined in the below table 18.

TABLE 18 2-bit field P_(I) N_(TI) 00 1 1 01 2 2 10 4 3 11 8 4

DP_FRAME_INTERVAL: This 2-bit field indicates the frame interval (I_(JUMP)) within the frame-group for the associated DP and the allowed values are 1, 2, 4, 8 (the corresponding 2-bit field is ‘00’, ‘01’, ‘10’, or ‘11’, respectively). For DPs that do not appear every frame of the frame-group, the value of this field is equal to the interval between successive frames. For example, if a DP appears on the frames 1, 5, 9, 13, etc., this field is set to ‘4’. For DPs that appear in every frame, this field is set to ‘1’.

DP_TI_BYPASS: This 1-bit field determines the availability of time interleaver. If time interleaving is not used for a DP, it is set to ‘1’. Whereas if time interleaving is used it is set to ‘0’.

DP_FIRST_FRAME_IDX: This 5-bit field indicates the index of the first frame of the super-frame in which the current DP occurs. The value of DP_FIRST_FRAME_IDX ranges from 0 to 31

DP_NUM_BLOCK_MAX: This 10-bit field indicates the maximum value of DP_NUM_BLOCKS for this DP. The value of this field has the same range as DP_NUM_BLOCKS.

DP_PAYLOAD_TYPE: This 2-bit field indicates the type of the payload data carried by the given DP. DP_PAYLOAD_TYPE is signaled according to the below table 19.

TABLE 19 Value Payload Type 00 TS. 01 IP 10 GS 11 reserved

DP_INBAND_MODE: This 2-bit field indicates whether the current DP carries in-band signaling information. The in-band signaling type is signaled according to the below table 20.

TABLE 20 Value In-band mode 00 In-band signaling is not carried. 01 INBAND-PLS is carried only 10 INBAND-ISSY is carried only 11 INBAND-PLS and INBAND-ISSY are carried

DP_PROTOCOL_TYPE: This 2-bit field indicates the protocol type of the payload carried by the given DP. It is signaled according to the below table 21 when input payload types are selected.

TABLE 21 If DP_PAYLOAD_TYPE If DP_PAYLOAD_TYPE If DP_PAYLOAD_TYPE Value Is TS Is IP Is GS 00 MPEG2-TS IPv4 (Note) 01 Reserved IPv6 Reserved 10 Reserved Reserved Reserved 11 Reserved Reserved Reserved

DP_CRC_MODE: This 2-bit field indicates whether CRC encoding is used in the Input Formatting block. The CRC mode is signaled according to the below table 22.

TABLE 22 Value CRC mode 00 Not used 01 CRC-8 10 CRC-16 11 CRC-32

DNP_MODE: This 2-bit field indicates the null-packet deletion mode used by the associated DP when DP_PAYLOAD_TYPE is set to TS (‘00’). DNP_MODE is signaled according to the below table 23. If DP_PAYLOAD_TYPE is not TS (‘00’), DNP_MODE is set to the value ‘00’.

TABLE 23 Value Null-packet deletion mode 00 Not used 01 DNP-NORMAL 10 DNP-OFFSET 11 reserved

ISSY_MODE: This 2-bit field indicates the ISSY mode used by the associated DP when DP_PAYLOAD_TYPE is set to TS (‘00’). The ISSY_MODE is signaled according to the below table 24 If DP_PAYLOAD_TYPE is not TS (‘00’), ISSY_MODE is set to the value ‘00’.

TABLE 24 Value ISSY mode 00 Not used 01 ISSY-UP 10 ISSY-BBF 11 reserved

HC_MODE_TS: This 2-bit field indicates the TS header compression mode used by the associated DP when DP_PAYLOAD_TYPE is set to TS (‘00’). The HC_MODE_TS is signaled according to the below table 25.

TABLE 25 Value Header compression mode 00 HC_MODE_TS 1 01 HC_MODE_TS 2 10 HC_MODE_TS 3 11 HC_MODE_TS 4

HC_MODE_IP: This 2-bit field indicates the IP header compression mode when DP_PAYLOAD_TYPE is set to IP (‘01’). The HC_MODE_IP is signaled according to the below table 26.

TABLE 26 Value Header compression mode 00 No compression 01 HC_MODE_IP 1 10~11 reserved

PID: This 13-bit field indicates the PID number for TS header compression when DP_PAYLOAD_TYPE is set to TS (‘00’) and HC_MODE_TS is set to ‘01’ or ‘10’.

RESERVED: This 8-bit field is reserved for future use.

The following field appears only if FIC_FLAG is equal to ‘1’:

FIC_VERSION: This 8-bit field indicates the version number of the FIC.

FIC_LENGTH_BYTE: This 13-bit field indicates the length, in bytes, of the FIC.

RESERVED: This 8-bit field is reserved for future use.

The following field appears only if AUX_FLAG is equal to ‘1’:

NUM_AUX: This 4-bit field indicates the number of auxiliary streams. Zero means no auxiliary streams are used.

AUX_CONFIG_RFU: This 8-bit field is reserved for future use.

AUX_STREAM_TYPE: This 4-bit is reserved for future use for indicating the type of the current auxiliary stream.

AUX_PRIVATE_CONFIG: This 28-bit field is reserved for future use for signaling auxiliary streams.

FIG. 15 illustrates PLS2 data according to another embodiment of the present invention.

FIG. 15 illustrates PLS2-DYN data of the PLS2 data. The values of the PLS2-DYN data may change during the duration of one frame-group, while the size of fields remains constant.

The details of fields of the PLS2-DYN data are as follows:

FRAME_INDEX: This 5-bit field indicates the frame index of the current frame within the super-frame. The index of the first frame of the super-frame is set to ‘0’.

PLS_CHANGE_COUNTER: This 4-bit field indicates the number of super-frames ahead where the configuration will change. The next super-frame with changes in the configuration is indicated by the value signaled within this field. If this field is set to the value ‘0000’, it means that no scheduled change is foreseen: e.g., value ‘1’ indicates that there is a change in the next super-frame.

FIC_CHANGE_COUNTER: This 4-bit field indicates the number of super-frames ahead where the configuration (i.e., the contents of the FIC) will change. The next super-frame with changes in the configuration is indicated by the value signaled within this field. If this field is set to the value ‘0000’, it means that no scheduled change is foreseen: e.g. value ‘0001’ indicates that there is a change in the next super-frame.

RESERVED: This 16-bit field is reserved for future use.

The following fields appear in the loop over NUM_DP, which describe the parameters associated with the DP carried in the current frame.

DP_ID: This 6-bit field indicates uniquely the DP within a PHY profile.

DP_START: This 15-bit (or 13-bit) field indicates the start position of the first of the DPs using the DPU addressing scheme. The DP_START field has differing length according to the PHY profile and FFT size as shown in the below table 27.

TABLE 27 DP_START field size PHY profile 64K 16K Base 13 bit 15 bit Handheld — 13 bit Advanced 13 bit 15 bit

DP_NUM_BLOCK: This 10-bit field indicates the number of FEC blocks in the current TI group for the current DP. The value of DP_NUM_BLOCK ranges from 0 to 1023

RESERVED: This 8-bit field is reserved for future use.

The following fields indicate the FIC parameters associated with the EAC.

EAC_FLAG: This 1-bit field indicates the existence of the EAC in the current frame. This bit is the same value as the EAC_FLAG in the preamble.

EAS_WAKE_UP_VERSION_NUM: This 8-bit field indicates the version number of a wake-up indication.

If the EAC_FLAG field is equal to ‘1’, the following 12 bits are allocated for EAC_LENGTH_BYTE field. If the EAC_FLAG field is equal to ‘0’, the following 12 bits are allocated for EAC_COUNTER.

EAC_LENGTH_BYTE: This 12-bit field indicates the length, in byte, of the EAC.

EAC_COUNTER: This 12-bit field indicates the number of the frames before the frame where the EAC arrives.

The following field appears only if the AUX_FLAG field is equal to ‘1’:

AUX_PRIVATE_DYN: This 48-bit field is reserved for future use for signaling auxiliary streams. The meaning of this field depends on the value of AUX_STREAM_TYPE in the configurable PLS2-STAT.

CRC_32: A 32-bit error detection code, which is applied to the entire PLS2.

FIG. 16 illustrates a logical structure of a frame according to an embodiment of the present invention.

As above mentioned, the PLS, EAC, FIC, DPs, auxiliary streams and dummy cells are mapped into the active carriers of the OFDM symbols in the frame. The PLS1 and PLS2 are first mapped into one or more FSS(s). After that, EAC cells, if any, are mapped immediately following the PLS field, followed next by FIC cells, if any. The DPs are mapped next after the PLS or EAC, FIC, if any. Type 1 DPs follows first, and Type 2 DPs next. The details of a type of the DP will be described later. In some case, DPs may carry some special data for EAS or service signaling data. The auxiliary stream or streams, if any, follow the DPs, which in turn are followed by dummy cells. Mapping them all together in the above mentioned order, i.e. PLS, EAC, FIC, DPs, auxiliary streams and dummy data cells exactly fill the cell capacity in the frame.

FIG. 17 illustrates PLS mapping according to an embodiment of the present invention.

PLS cells are mapped to the active carriers of FSS(s). Depending on the number of cells occupied by PLS, one or more symbols are designated as FSS(s), and the number of FSS(s) N_(FSS) is signaled by NUM_FSS in PLS1. The FSS is a special symbol for carrying PLS cells. Since robustness and latency are critical issues in the PLS, the FSS(s) has higher density of pilots allowing fast synchronization and frequency-only interpolation within the FSS.

PLS cells are mapped to active carriers of the N_(FSS) FSS(s) in a top-down manner as shown in an example in FIG. 17. The PLS1 cells are mapped first from the first cell of the first FSS in an increasing order of the cell index. The PLS2 cells follow immediately after the last cell of the PLS1 and mapping continues downward until the last cell index of the first FSS. If the total number of required PLS cells exceeds the number of active carriers of one FSS, mapping proceeds to the next FSS and continues in exactly the same manner as the first FSS.

After PLS mapping is completed, DPs are carried next. If EAC, FIC or both are present in the current frame, they are placed between PLS and “normal” DPs.

FIG. 18 illustrates EAC mapping according to an embodiment of the present invention.

EAC is a dedicated channel for carrying EAS messages and links to the DPs for EAS. EAS support is provided but EAC itself may or may not be present in every frame. EAC, if any, is mapped immediately after the PLS2 cells. EAC is not preceded by any of the FIC, DPs, auxiliary streams or dummy cells other than the PLS cells. The procedure of mapping the EAC cells is exactly the same as that of the PLS.

The EAC cells are mapped from the next cell of the PLS2 in increasing order of the cell index as shown in the example in FIG. 18. Depending on the EAS message size, EAC cells may occupy a few symbols, as shown in FIG. 18.

EAC cells follow immediately after the last cell of the PLS2, and mapping continues downward until the last cell index of the last FSS. If the total number of required EAC cells exceeds the number of remaining active carriers of the last FSS mapping proceeds to the next symbol and continues in exactly the same manner as FSS(s). The next symbol for mapping in this case is the normal data symbol, which has more active carriers than a FSS.

After EAC mapping is completed, the FIC is carried next, if any exists. If FIC is not transmitted (as signaled in the PLS2 field), DPs follow immediately after the last cell of the EAC.

FIG. 19 illustrates FIC mapping according to an embodiment of the present invention.

(a) shows an example mapping of FIC cell without EAC and (b) shows an example mapping of FIC cell with EAC.

FIC is a dedicated channel for carrying cross-layer information to enable fast service acquisition and channel scanning. This information primarily includes channel binding information between DPs and the services of each broadcaster. For fast scan, a receiver can decode FIC and obtain information such as broadcaster ID, number of services, and BASE_DP_ID. For fast service acquisition, in addition to FIC, base DP can be decoded using BASE_DP_ID. Other than the content it carries, a base DP is encoded and mapped to a frame in exactly the same way as a normal DP. Therefore, no additional description is required for a base DP. The FIC data is generated and consumed in the Management Layer. The content of FIC data is as described in the Management Layer specification.

The FIC data is optional and the use of FIC is signaled by the FIC_FLAG parameter in the static part of the PLS2. If FIC is used, FIC_FLAG is set to ‘1’ and the signaling field for FIC is defined in the static part of PLS2. Signaled in this field are FIC_VERSION, and FIC_LENGTH_BYTE. FIC uses the same modulation, coding and time interleaving parameters as PLS2. FIC shares the same signaling parameters such as PLS2_MOD and PLS2_FEC. FIC data, if any, is mapped immediately after PLS2 or EAC if any. FIC is not preceded by any normal DPs, auxiliary streams or dummy cells. The method of mapping FIC cells is exactly the same as that of EAC which is again the same as PLS.

Without EAC after PLS, FIC cells are mapped from the next cell of the PLS2 in an increasing order of the cell index as shown in an example in (a). Depending on the FIC data size, FIC cells may be mapped over a few symbols, as shown in (b).

FIC cells follow immediately after the last cell of the PLS2, and mapping continues downward until the last cell index of the last FSS. If the total number of required FIC cells exceeds the number of remaining active carriers of the last FSS, mapping proceeds to the next symbol and continues in exactly the same manner as FSS(s). The next symbol for mapping in this case is the normal data symbol which has more active carriers than a FSS.

If EAS messages are transmitted in the current frame, EAC precedes FIC, and FIC cells are mapped from the next cell of the EAC in an increasing order of the cell index as shown in (b).

After FIC mapping is completed, one or more DPs are mapped, followed by auxiliary streams, if any, and dummy cells.

FIG. 20 illustrates a type of DP according to an embodiment of the present invention.

(a) shows type 1 DP and (b) shows type 2 DP.

After the preceding channels, i.e., PLS, EAC and FIC, are mapped, cells of the DPs are mapped. A DP is categorized into one of two types according to mapping method:

Type 1 DP: DP is mapped by TDM

Type 2 DP: DP is mapped by FDM

The type of DP is indicated by DP_TYPE field in the static part of PLS2. FIG. 20 illustrates the mapping orders of Type 1 DPs and Type 2 DPs. Type 1 DPs are first mapped in the increasing order of cell index, and then after reaching the last cell index, the symbol index is increased by one. Within the next symbol, the DP continues to be mapped in the increasing order of cell index starting from p=0. With a number of DPs mapped together in one frame, each of the Type 1 DPs are grouped in time, similar to TDM multiplexing of DPs.

Type 2 DPs are first mapped in the increasing order of symbol index, and then after reaching the last OFDM symbol of the frame, the cell index increases by one and the symbol index rolls back to the first available symbol and then increases from that symbol index. After mapping a number of DPs together in one frame, each of the Type 2 DPs are grouped in frequency together, similar to FDM multiplexing of DPs.

Type 1 DPs and Type 2 DPs can coexist in a frame if needed with one restriction; Type 1 DPs always precede Type 2 DPs. The total number of OFDM cells carrying Type 1 and Type 2 DPs cannot exceed the total number of OFDM cells available for transmission of DPs:

D _(DP1) +D _(DP2) ≦DP  [Expression 2]

where D_(DP1) is the number of OFDM cells occupied by Type 1 DPs, D_(DP2) is the number of cells occupied by Type 2 DPs. Since PLS, EAC, FIC are all mapped in the same way as Type 1 DP, they all follow “Type 1 mapping rule”. Hence, overall, Type 1 mapping always precedes Type 2 mapping.

FIG. 21 illustrates DP mapping according to an embodiment of the present invention.

(a) shows an addressing of OFDM cells for mapping type 1 DPs and (b) shows an an addressing of OFDM cells for mapping for type 2 DPs.

Addressing of OFDM cells for mapping Type 1 DPs (0, . . . , D_(DP1)−1) is defined for the active data cells of Type 1 DPs. The addressing scheme defines the order in which the cells from the TIs for each of the Type 1 DPs are allocated to the active data cells. It is also used to signal the locations of the DPs in the dynamic part of the PLS2.

Without EAC and FIC, address 0 refers to the cell immediately following the last cell carrying PLS in the last FSS. If EAC is transmitted and FIC is not in the corresponding frame, address 0 refers to the cell immediately following the last cell carrying EAC. If FIC is transmitted in the corresponding frame, address 0 refers to the cell immediately following the last cell carrying FIC. Address 0 for Type 1 DPs can be calculated considering two different cases as shown in (a). In the example in (a), PLS, EAC and FIC are assumed to be all transmitted. Extension to the cases where either or both of EAC and FIC are omitted is straightforward. If there are remaining cells in the FSS after mapping all the cells up to FIC as shown on the left side of (a).

Addressing of OFDM cells for mapping Type 2 DPs (0, . . . , D_(DP2)−1) is defined for the active data cells of Type 2 DPs. The addressing scheme defines the order in which the cells from the TIs for each of the Type 2 DPs are allocated to the active data cells. It is also used to signal the locations of the DPs in the dynamic part of the PLS2.

Three slightly different cases are possible as shown in (b). For the first case shown on the left side of (b), cells in the last FSS are available for Type 2 DP mapping. For the second case shown in the middle, FIC occupies cells of a normal symbol, but the number of FIC cells on that symbol is not larger than C_(FSS). The third case, shown on the right side in (b), is the same as the second case except that the number of FIC cells mapped on that symbol exceeds C_(FSS).

The extension to the case where Type 1 DP(s) precede Type 2 DP(s) is straightforward since PLS, EAC and FIC follow the same “Type 1 mapping rule” as the Type 1 DP(s).

A data pipe unit (DPU) is a basic unit for allocating data cells to a DP in a frame.

A DPU is defined as a signaling unit for locating DPs in a frame. A Cell Mapper 7010 may map the cells produced by the TIs for each of the DPs. A Time interleaver 5050 outputs a series of TI-blocks and each TI-block comprises a variable number of XFECBLOCKs which is in turn composed of a set of cells. The number of cells in an XFECBLOCK, N_(cells), is dependent on the FECBLOCK size, N_(ldpc), and the number of transmitted bits per constellation symbol. A DPU is defined as the greatest common divisor of all possible values of the number of cells in a XFECBLOCK, N_(cells), supported in a given PHY profile. The length of a DPU in cells is defined as L_(DPU). Since each PHY profile supports different combinations of FECBLOCK size and a different number of bits per constellation symbol, L_(DPU) is defined on a PHY profile basis.

FIG. 22 illustrates an FEC structure according to an embodiment of the present invention.

FIG. 22 illustrates an FEC structure according to an embodiment of the present invention before bit interleaving. As above mentioned, Data FEC encoder may perform the FEC encoding on the input BBF to generate FECBLOCK procedure using outer coding (BCH), and inner coding (LDPC). The illustrated FEC structure corresponds to the FECBLOCK. Also, the FECBLOCK and the FEC structure have same value corresponding to a length of LDPC codeword.

The BCH encoding is applied to each BBF (K_(bch) bits), and then LDPC encoding is applied to BCH-encoded BBF (K_(ldpc) bits=N_(bch) bits) as illustrated in FIG. 22.

The value of N_(ldpc) is either 64800 bits (long FECBLOCK) or 16200 bits (short FECBLOCK).

The below table 28 and table 29 show FEC encoding parameters for a long FECBLOCK and a short FECBLOCK, respectively.

TABLE 28 BCH error LDPC correction Rate Nldpc Kldpc Kbch capability Nbch-Kbch  5/15 64800 21600 21408 12 192  6/15 25920 25728  7/15 30240 30048  8/15 34560 34368  9/15 38880 38688 10/15 43200 43008 11/15 47520 47328 12/15 51840 51648 13/15 56160 55968

TABLE 29 BCH error LDPC correction Rate N_(ldpc) K_(ldpc) K_(bch) capability N_(bch)-K_(bch)  5/15 16200 5400 5232 12 168  6/15 6480 6312  7/15 7560 7392  8/15 8640 8472  9/15 9720 9552 10/15 10800 10632 11/15 11880 11712 12/15 12960 12792 13/15 14040 13872

The details of operations of the BCH encoding and LDPC encoding are as follows:

A 12-error correcting BCH code is used for outer encoding of the BBF. The BCH generator polynomial for short FECBLOCK and long FECBLOCK are obtained by multiplying together all polynomials.

LDPC code is used to encode the output of the outer BCH encoding. To generate a completed B_(ldpc) (FECBLOCK), P_(ldpc) (parity bits) is encoded systematically from each I_(ldpc) (BCH-encoded BBF), and appended to I_(ldpc). The completed B_(ldpc) (FECBLOCK) are expressed as follow Expression.

B _(ldpc) =[I _(ldpc) P _(ldpc) ]=[i ₀ ,i ₁ , . . . ,i _(K) _(ldpc) ⁻¹ ,p ₀ ,p ₁ , . . . ,p _(N) _(ldpc) _(−K) _(ldpc) ⁻¹]  [Expression 3]

The parameters for long FECBLOCK and short FECBLOCK are given in the above table 28 and 29, respectively.

The detailed procedure to calculate N_(ldpc)−K_(ldpc) parity bits for long FECBLOCK, is as follows:

1) Initialize the parity bits,

p ₀ =p ₁ =p ₂ = . . . =p _(N) _(ldpc) _(−K) _(ldpc) ⁻¹=0  [Expression 4]

2) Accumulate the first information bit—i₀, at parity bit addresses specified in the first row of an addresses of parity check matrix. The details of addresses of parity check matrix will be described later. For example, for rate 13/15:

p ₉₈₃ =p ₉₈₃ ⊕i ₀ p ₂₈₁₅ =p ₂₈₁₅ ⊕i ₀

p ₄₈₃₇ =p ₄₈₃₇ ⊕i ₀ p ₄₉₈₉ =p ₄₉₈₉ ⊕i ₀

p ₆₁₃₈ =p ₆₁₃₃ ⊕i ₀ p ₆₄₅₈ =p ₆₄₅₃ ⊕i ₀

p ₆₉₂₁ =p ₆₉₂₁ ⊕i ₀ p ₆₉₇₄ =p ₆₉₇₄ ⊕i ₀

p ₇₅₇₂ =p ₇₅₇₂ ⊕i ₀ p ₈₂₆₀ =p ₈₂₆₀ ⊕i ₀

p ₈₄₉₆ =p ₈₄₉₆ ⊕i ₀  [Expression 5]

3) For the next 359 information bits, i_(s), s=1, 2, . . . , 359 accumulate i_(s) at parity bit addresses using following Expression.

{x+(s mod 360)=Q _(ldpc)} mod(N _(ldpc) −K _(ldpc))  [Expression 6]

where x denotes the address of the parity bit accumulator corresponding to the first bit i₀, and Q_(ldpc) is a code rate dependent constant specified in the addresses of parity check matrix. Continuing with the example, Q_(ldpc)=24 for rate 13/15, so for information bit i₁, the following operations are performed:

p ₁₀₀₇ =p ₁₀₀₇ ⊕i ₁ p ₂₈₃₉ =p ₂₈₃₉ ⊕i ₁

p ₄₈₆₁ =p ₄₈₆₁ ⊕i ₁ p ₅₀₁₃ =p ₅₀₁₃ ⊕i ₁

p ₆₁₆₂ =p ₆₁₆₂ ⊕i ₁ p ₆₄₈₂ =p ₆₄₈₂ ⊕i ₁

p ₆₉₄₅ =p ₆₉₄₅ ⊕i ₁ p ₆₉₉₈ =p ₆₉₉₈ ⊕i ₁

p ₇₅₉₆ =p ₇₅₉₆ ⊕i ₁ p ₈₇₈₄ =p ₈₂₈₄ ⊕i ₁

p ₈₅₂₀ =p ₈₅₂₀ ⊕i ₁  [Expression 7]

4) For the 361^(st) information bit i₃₆₀, the addresses of the parity bit accumulators are given in the second row of the addresses of parity check matrix. In a similar manner the addresses of the parity bit accumulators for the following 359 information bits i_(s), s=361, 362, . . . , 719 are obtained using the Expression 6, where x denotes the address of the parity bit accumulator corresponding to the information bit i₃₆₀, i.e., the entries in the second row of the addresses of parity check matrix.

5) In a similar manner, for every group of 360 new information bits, a new row from addresses of parity check matrixes used to find the addresses of the parity bit accumulators.

After all of the information bits are exhausted, the final parity bits are obtained as follows:

6) Sequentially perform the following operations starting with i=1

p _(i) =p _(i) ⊕p _(i-1) ,i=1,2, . . . ,N _(ldpc) −K _(ldpc)−1  [Expression 8]

where final content of p_(i), i=0, 1, . . . N_(ldpc)−K_(ldpc)−1 is equal to the parity bit p₁.

TABLE 30 Code Rate Q_(ldpc)  5/15 120  6/15 108  7/15 96  8/15 84  9/15 72 10/15 60 11/15 48 12/15 36 13/15 24

This LDPC encoding procedure for a short FECBLOCK is in accordance with t LDPC encoding procedure for the long FECBLOCK, except replacing the table 30 with table 31, and replacing the addresses of parity check matrix for the long FECBLOCK with the addresses of parity check matrix for the short FECBLOCK.

TABLE 31 Code Rate Q_(ldpc)  5/15 30  6/15 27  7/15 24  8/15 21  9/15 18 10/15 15 11/15 12 12/15 9 13/15 6

FIG. 23 illustrates a bit interleaving according to an embodiment of the present invention.

The outputs of the LDPC encoder are bit-interleaved, which consists of parity interleaving followed by Quasi-Cyclic Block (QCB) interleaving and inner-group interleaving.

(a) shows Quasi-Cyclic Block (QCB) interleaving and (b) shows inner-group interleaving.

The FECBLOCK may be parity interleaved. At the output of the parity interleaving, the LDPC codeword consists of 180 adjacent QC blocks in a long FECBLOCK and 45 adjacent QC blocks in a short FECBLOCK. Each QC block in either a long or short FECBLOCK consists of 360 bits. The parity interleaved LDPC codeword is interleaved by QCB interleaving. The unit of QCB interleaving is a QC block. The QC blocks at the output of parity interleaving are permutated by QCB interleaving as illustrated in FIG. 23, where N_(cells)=64800/η_(mod) or 16200/η_(mod) according to the FECBLOCK length. The QCB interleaving pattern is unique to each combination of modulation type and LDPC code rate.

After QCB interleaving, inner-group interleaving is performed according to modulation type and order (η_(mod)) which is defined in the below table 32. The number of QC blocks for one inner-group, N_(QCB) _(_) _(IG), is also defined.

TABLE 32 Modulation type η_(mod) N_(QCB)_IG QAM-16 4 2 NUC-16 4 4 NUQ-64 6 3 NUC-64 6 6 NUQ-256 8 4 NUC-256 8 8 NUQ-1024 10 5 NUC-1024 10 10

The inner-group interleaving process is performed with N_(QCB) _(_) _(IG) QC blocks of the QCB interleaving output. Inner-group interleaving has a process of writing and reading the bits of the inner-group using 360 columns and N_(QCB) _(_) _(IG) rows. In the write operation, the bits from the QCB interleaving output are written row-wise. The read operation is performed column-wise to read out m bits from each row, where m is equal to 1 for NUC and 2 for NUQ.

FIG. 24 illustrates a cell-word demultiplexing according to an embodiment of the present invention.

(a) shows a cell-word demultiplexing for 8 and 12 bpcu MIMO and (b) shows a cell-word demultiplexing for 10 bpcu MIMO.

Each cell word (c_(0,1), c_(1,1), . . . , c_(ηmod-1,1)) of the bit interleaving output is demultiplexed into (d_(1,0,m), d_(1,1,m) . . . , d_(1,ηmod-1,m)) and (d_(2,0,m), d_(2,1,m) . . . , d_(2,ηmod-1,m)) as shown in (a), which describes the cell-word demultiplexing process for one XFECBLOCK.

For the 10 bpcu MIMO case using different types of NUQ for MIMO encoding, the Bit Interleaver for NUQ-1024 is re-used. Each cell word (c_(0,1), c_(1,1), . . . , c_(9,1)) of the Bit Interleaver output is demultiplexed into (d_(1,0,m), d_(1,1,m) . . . , d_(1,3,m)) and (d_(2,0,m), d_(2,1,m) . . . , d_(2,5,m)), as shown in (b).

FIG. 25 illustrates a time interleaving according to an embodiment of the present invention.

(a) to (c) show examples of TI mode.

The time interleaver operates at the DP level. The parameters of time interleaving (TI) may be set differently for each DP.

The following parameters, which appear in part of the PLS2-STAT data, configure the TI:

DP_TI_TYPE (allowed values: 0 or 1): Represents the TI mode; ‘0’ indicates the mode with multiple TI blocks (more than one TI block) per TI group. In this case, one TI group is directly mapped to one frame (no inter-frame interleaving). ‘1’ indicates the mode with only one TI block per TI group. In this case, the TI block may be spread over more than one frame (inter-frame interleaving).

DP_TI_LENGTH: If DP_TI_TYPE=‘0’, this parameter is the number of TI blocks N_(TI) per TI group. For DP_TI_TYPE=‘1’, this parameter is the number of frames P_(I) spread from one TI group.

DP_NUM_BLOCK_MAX (allowed values: 0 to 1023): Represents the maximum number of XFECBLOCKs per TI group.

DP_FRAME_INTERVAL (allowed values: 1, 2, 4, 8): Represents the number of the frames L_(JUMP) between two successive frames carrying the same DP of a given PHY profile.

DP_TI_BYPASS (allowed values: 0 or 1): If time interleaving is not used for a DP, this parameter is set to ‘1’. It is set to ‘0’ if time interleaving is used.

Additionally, the parameter DP_NUM_BLOCK from the PLS2-DYN data is used to represent the number of XFECBLOCKs carried by one TI group of the DP.

When time interleaving is not used for a DP, the following TI group, time interleaving operation, and TI mode are not considered. However, the Delay Compensation block for the dynamic configuration information from the scheduler will still be required. In each DP, the XFECBLOCKs received from the SSD/MIMO encoding are grouped into TI groups. That is, each TI group is a set of an integer number of XFECBLOCKs and will contain a dynamically variable number of XFECBLOCKs. The number of XFECBLOCKs in the TI group of index n is denoted by N_(xBLock) _(_) _(Group)(n) and is signaled as DP_NUM_BLOCK in the PLS2-DYN data. Note that N_(xBLOCK) _(_) _(Group)(n) may vary from the minimum value of 0 to the maximum value N_(xBLocx) _(_) _(Group) _(_) _(MAX) (corresponding to DP_NUM_BLOCK_MAX) of which the largest value is 1023.

Each TI group is either mapped directly onto one frame or spread over P_(I) frames. Each TI group is also divided into more than one TI blocks(N_(TI)), where each TI block corresponds to one usage of time interleaver memory. The TI blocks within the TI group may contain slightly different numbers of XFECBLOCKs. If the TI group is divided into multiple TI blocks, it is directly mapped to only one frame. There are three options for time interleaving (except the extra option of skipping the time interleaving) as shown in the below table 33.

TABLE 33 Modes Descriptions Option-1 Each TI group contains one TI block and is mapped directly to one frame as shown in (a). This option is signaled in the PLS2-STAT by DP_TI_TYPE = ″0″ and DP_TI_ LENGTH = ′1′(N_(TI) = 1). Option-2 Each TI group contains one TI block and is mapped to more than one frame. (b) shows an example, where one TI group is mapped to two frames, i.e., DP_TI_LENGTH = ′2′ (P_(I) = 2) and DP_FRAME_INTERVAL (I_(JUMP) = 2). This provides greater time diversity for low data-rate services. This option is signaled in the PLS2-STAT by DP_TI_ TYPE = ′1′. Option-3 Each TI group is divided into multiple TI blocks and is mapped directly to one frame as shown in (c). Each TI block may use full TI memory, so as to provide the maximum bit-rate for a DP. This option is signaled in the PLS2-STAT signaling by DP_TI_TYPE = ′0′ and DP_TI_LENGTH = N_(TI), while P_(I) = 1.

In each DP, the TI memory stores the input XFECBLOCKs (output XFECBLOCKs from the SSD/MIMO encoding block). Assume that input XFECBLOCKs are defined as

$d_{n,s,r,q} = \left\{ \begin{matrix} {f_{n,s,r,q},} & {{theoutputof}\mspace{14mu} {SSD}\mspace{14mu} \ldots \mspace{14mu} {encoding}} \\ {g_{n,s,r,q},} & {{theoutputof}\mspace{14mu} {MIMOencoding}} \end{matrix} \right.$

where d_(n,s,r,q) is the qth cell of the rth XFECBLOCK in the sth TI block of the nth TI group and represents the outputs of SSD and MIMO encodings as follows.

(d_(n, s, 0, 0), d_(n, s, 0, 1), …  , d_(n, s, 0, N_(cells) − 1), d_(n, s, 1, 0), …  , d_(n, s, 1, N_(cells) − 1), …  , d_(n, s, N_(xBLOCK_TI)(n, s)), …  , d_(n, s, N_(xBLOCK_TI)(n, s) − 1, N_(cells) − 1)),

In addition, assume that output XFECBLOCKs from the time interleaver are defined as

(h_(n, s, 0), h_(n, s, 1), …  , h_(n, s, i), …  , h_(n, s, N_(xBLOCK  TI)(n, s) × N_(cells) − 1))

where n_(n,s,t) is the ith output cell (for i=0, . . . , N_(xBLOCK) _(_) _(TI)(n,s)×N_(cells)−1) in the sth TI block of the nth TI group.

Typically, the time interleaver will also act as a buffer for DP data prior to the process of frame building. This is achieved by means of two memory banks for each DP. The first TI-block is written to the first bank. The second TI-block is written to the second bank while the first bank is being read from and so on.

The TI is a twisted row-column block interleaver. For the sth TI block of the nth TI group, the number of rows N_(r) of a TI memory is equal to the number of cells N_(cells), i.e., N_(r)=N_(cell), while the number of columns N_(c) is equal to the number N_(xBLOCK) _(_) _(TI)(n,s).

FIG. 26 illustrates the basic operation of a twisted row-column block interleaver according to an embodiment of the present invention.

shows a writing operation in the time interleaver and (b) shows a reading operation in the time interleaver The first XFECBLOCK is written column-wise into the first column of the TI memory, and the second XFECBLOCK is written into the next column, and so on as shown in (a). Then, in the interleaving array, cells are read out diagonal-wise. During diagonal-wise reading from the first row (rightwards along the row beginning with the left-most column) to the last row, N_(r) cells are read out as shown in (b). In detail, assuming z_(n,s,i)(i=0, . . . , N_(r)N_(c)) as the TI memory cell position to be read sequentially, the reading process in such an interleaving array is performed by calculating the row index R_(n,s,i), the column index C_(n,s,i), and the associated twisting parameter as follows expression.

$\begin{matrix} {{{GENERATE}\left( {R_{n,s,i},C_{n,s,i}} \right)} = \left\{ {{R_{n,s,i} = {{mod}\left( {i,N_{r}} \right)}},{T_{n,s,i} = {{mod}\left( {{S_{shift} \times R_{n,s,i}},N_{c}} \right)}},{C_{n,s,i} = {{mod}\left( {{T_{n,s,i} + \left\lfloor \frac{i}{N_{r}} \right\rfloor},N_{c}} \right)}}} \right\}} & \left\lbrack {{Expression}\mspace{14mu} 9} \right\rbrack \end{matrix}$

where S_(shift) is a common shift value for the diagonal-wise reading process regardless of N_(xBLOCK) _(_) _(TI)(n,s), and it is determined by N_(xBLOCK) _(_) _(TI) _(_) _(MAX) given in the PLS2-STAT as follows expression.

$\begin{matrix} {{for}\left\{ {\begin{matrix} {N_{{xBLOCK\_ TI}{\_ MAX}}^{\prime} =} & {if} \\ {{N_{{xBLOCK\_ TI}{\_ MAX}} + 1},} & {{N_{{xBLOCK\_ TI}{\_ MAX}}{mod}\; 2} = 0} \\ {N_{{xBLOCK}\mspace{14mu} {TI}\mspace{14mu} {MAX}}^{\prime} =} & {if} \\ {N_{{xBLOCK}\mspace{14mu} {TI}\mspace{14mu} {MAX}},} & {{N_{{xBLOCK}\mspace{14mu} {TI}\mspace{14mu} {MAX}}{mod}\; 2} = 1} \end{matrix},{S_{shift} = \frac{N_{{xBLOCK\_ TI}{\_ MAX}}^{\prime} - 1}{2}}} \right.} & \left\lbrack {{Expression}\mspace{14mu} 10} \right\rbrack \end{matrix}$

As a result, the cell positions to be read are calculated by a coordinate as z_(n,s,i)=N_(r)C_(n,s,i)+R_(n,s,i).

FIG. 27 illustrates an operation of a twisted row-column block interleaver according to another embodiment of the present invention.

More specifically, FIG. 27 illustrates the interleaving array in the TI memory for each TI group, including virtual XFECBLOCKs when N_(xBLOCK) _(_) _(TI)(0,0)=3, N_(xBLOCK) _(_) _(TI)(1,0)=6, N_(xBLOCK) _(_) _(TI)(2,0)=5.

The variable number N_(xBLOCK) _(_) _(TI)(n,s)=N_(r) will be less than or equal to N′_(xBLOCK) _(_) _(TI) _(_) _(MAX). Thus, in order to achieve a single-memory deinterleaving at the receiver side, regardless of N_(xBLOCK TI)(n,s), the interleaving array for use in a twisted row-column block interleaver is set to the size of N_(r)×N_(c)=N_(cell)×N′_(xBLOCK) _(_) _(TI) _(_) _(MAX) by inserting the virtual XFECBLOCKs into the TI memory and the reading process is accomplished as follow expression.

 

 Expression 11 

p = 0; for i = 0; i < N_(cells)N′_(χBLOCK)_TI_MAX; i = i + 1 {GENERATE (R_(n,s,i), C_(n,s,i)); V_(i) = N_(r)C_(n,s,j) + R_(n,s,j)  if V_(i) < N_(cells)N_(χBLOCK)_TI(n,s)  {   Z_(n,s,p) = V_(i); p = p + 1;   } }

The number of TI groups is set to 3. The option of time interleaver is signaled in the PLS2-STAT data by DP_TI_TYPE=‘0’, DP_FRAME_INTERVAL=‘1’, and DP_TI_LENGTH=‘1’, I_(JUMP)=1, and P_(I)=1. The number of XFECBLOCKs, each of which has N_(cells)=30 cells, per TI group is signaled in the PLS2-DYN data by N_(xBLOCK) _(_) _(TI)(0,0)=3, N_(xBLOCK) _(_) _(TI)(1,0)=6, and N_(xBLOCK) _(_) _(TI)(2,0)=5, respectively. The maximum number of XFECBLOCK is signaled in the PLS2-STAT data by N_(xBLOCK) _(_) _(Group) _(_) _(MAX), which leads to └N_(xBLOCK) _(_) _(Group) _(_) _(MAX)/N_(TI)┘=N_(xBLOCK) _(_) _(TI) _(_) _(MAX)=6.

FIG. 28 illustrates a diagonal-wise reading pattern of a twisted row-column block interleaver according to an embodiment of the present invention.

More specifically FIG. 28 shows a diagonal-wise reading pattern from each interleaving array with parameters of N′_(xBLOCK) _(_) _(TI) _(_) _(MAX)=7 and S_(shift)=(7−1)/2=3. Note that in the reading process shown as pseudocode above, if V_(i)≧N_(cells)N_(xBLOCK) _(_) _(TI)(n,s), the value of V_(i) is skipped and the next calculated value of V_(i) is used.

FIG. 29 illustrates interleaved XFECBLOCKs from each interleaving array according to an embodiment of the present invention.

FIG. 29 illustrates the interleaved XFECBLOCKs from each interleaving array with parameters of N′_(xBLOCK) _(_) _(TI) _(_) _(MAX)=7 and S_(shift)=3.

FIG. 30 shows a parity check matrix of a QC-IRA (quasi-cyclic irregular repeat accumulate) LDPC code.

The above-described LDPC encoder may encode a parity of an LDPC encoding block using the parity check matrix.

The parity check matrix according to the present invention is a parity check matrix of the QC-IRA LDPC code and may have the form of a quasi-cyclic matrix called an H matrix and be represented as H_(qc).

(a) shows a parity check matrix according to an embodiment of the present invention. As shown in (a), the parity check matrix is a matrix having a horizontal size of Q×(K+M) and a vertical size of Q×M and may include an information part and a parity part. The information part may include a matrix having a horizontal size of Q×K and a vertical size of Q×M and the parity part may include a matrix having a horizontal size of Q×M and a vertical size of Q×M. In this case, an LDPC code rate corresponds to K/(K+M).

The parity check matrix according to an embodiment of the present invention may include randomly distributed 1s and 0s and 1 may be referred to as an “edge”. The position of 1 in the parity check matrix, that is, the position of each edge may be represented in the form of a circulant shifted identity matrix per submatrix having a horizontal size of Q and a vertical size of Q. That is, a submatrix can be represented as a Q×Q circulant-shifted identity matrix including only 1 and 0. Specifically, the submatrix according to an embodiment of the present invention is represented as identity matrices I_(x) including I₀, I₁, I₂, {hacek over ( )}I₁ . . . , which have different positions of 1s according to the number of circulant shifts. The number of submatrices according to an embodiment of the present invention may be (K+M)×M.

(b) shows the circulant-shifted identity matrices I_(x) which represent submatrices according to an embodiment of the present invention.

The subscript x of I_(x) indicates the number of left-circulant shifts of columns of a circulant-shifted identity matrix. That is, I₁ represents an identify matrix in which columns are circulant-shifted to the left once and I₂ represents an identity matrix in which columns are circulant-shifted to the left twice. I_(Q) which is an identity matrix circulant-shifted Q times corresponding to the total number of columns, Q, may be the same matrix as I₀ due to circulant characteristic thereof.

I₀₊₂ represents a submatrix corresponding to a combination of two circulant-shifted identity matrices. In this case, the submatrix corresponds to a combination of the identity matrix I₀ and an identity matrix circulant-shifted twice.

{hacek over ( )}I₁ represents a circulant-shifted identity matrix in which the edge of the last column of the corresponding submatrix, that is, 1 has been removed while corresponding to the submatrix I₁.

The parity part of the parity check matrix of the QC-IRA LDPC code may include only submatrices I₀ and {hacek over ( )}I₁ and the position of submatrices I₀ may be fixed. As shown in (a), submatrices I₀ may be distributed in a diagonal direction in the parity part.

An edge in the parity check matrix represents that the corresponding row (checksum node) and the corresponding column (variable node) are physically connected. In this case, the number of 1s included in each column (variable node) may be referred to as a degree and columns may have the same degree or different degrees. Accordingly, the number, positions and value x of identity matrices I_(x) that represent edges grouped per submatrix are important factors in determining QC-IRA LDPC encoding performance and unique values may be determined per code rate.

(c) shows a base matrix of the parity check matrix according to an embodiment of the present invention. The base matrix represents only the number and positions of identity matrices I_(x) as specific numbers, ignoring the value x of I_(x). As shown in (c), a base matrix may have a horizontal size of K+M and a vertical size of M and be represented as H_(base). When I_(x) is not a matrix corresponding to a combination of submatrices, the position of the corresponding submatrix may be represented as 1. When a submatrix is represented as I₀₊₂, this submatrix is a matrix corresponding to a combination of two circulant-shifted identity matrices and thus the submatrix needs to be discriminated from a submatrix represented as one circulant-shifted identity matrix. In this case, the position of the submatrix may be represented as 2 which is the number of the combined circulant-shifted identity matrices. In the same manner, the position of a submatrix corresponding to a combination of N circulant-shifted identity matrices can be represented as N.

FIG. 31 shows a process of encoding the QC-IRA LDPC code according to an embodiment of the present invention.

The QC-IRA LDPC code may be encoded per submatrix, distinguished from conventional sequential encoding, to reduce processing complexity.

(a) shows arrangement of a QC-IRA parity check matrix in a QC form. The QC-IRA parity check matrix may be divided into 6 regions A, B, C, D, E and T when arranged in the QC form. When a Q×K information vector s, a parity vector p1 having a length of Q and a parity vector p2 having a length of Q×(M−1) are used, a codeword x can be represented as x={s, p1, p2}.

When the efficient encoding equation of Richardson is used, the codeword x can be obtained by directly acquiring p1 and p2 from the parity check matrix arranged in the QC form. The efficient encoding equation of Richardson is as follows.

φ=−ET ⁻¹ B+D

p ₁ ^(T)=−φ⁻¹(−ET ⁻¹ A+C)s ^(T)

p ₂ ^(T) =−T ⁻¹(As ^(T) +Bp ₁ ^(T))  [Expression 12]

(b) shows matrices φ and φ⁻¹ derived according to the efficient encoding equation.

As shown in (b), φ⁻¹ can be represented as a left low triangular (sub) matrix. The parity vector p2 can be obtained by calculating s and p1 according to the above-described equation. When the QC-IRA parity check matrix is encoded according to the efficient encoding equation of Richardson, at least Q parity nodes can be simultaneously processed in parallel according to characteristics of a Q×Q submatrix.

FIGS. 32 to 35 illustrate a process of sequentially encoding the QC-IRA LDPC code according to an embodiment of the present invention. This sequentially encoding may correspond to the above mentioned LDPC encoding.

FIG. 32 illustrates a parity check matrix permutation process according to an embodiment of the present invention.

(a) shows a QC-IRA LDPC parity check matrix H₁ arranged in QC form. As shown in (a), a parity part of the matrix H₁ may include submatrices distributed in a stepped form, which corresponds to the above-described QC-IRA LDPC parity check matrix. To easily perform sequential encoding, rows and columns of the matrix H₁ are moved such that the matrix H₁ is modified into a matrix H₂ according to an embodiment of the present invention.

(b) shows the modified matrix H₂. As shown in (b), a parity part of the matrix H₂ may include a dual diagonal matrix. In this case, an applied row and column permutation expression is as follows.

r _(y)=(r _(x) mod Q)M+└r _(x) /Q┘ where r _(x)=0,1,2, . . . ,QM−1

c _(y)={((c _(x) −QK)mod Q}M+└(c _(x) −QK/Q┘+QK where c _(x) =QK,QK+1, . . . ,Q(K+M)−1  [Expression 13]

According to the above permutation equation, the r_(x)-th row of the matrix H₁ can be moved to the r_(y)-th row of the matrix H₂ and the c_(x)-th column of the matrix H₁ can be moved to the c_(y)-th column of the matrix H₂. In this case, column permutation can be applied only to a parity processing period (QK≦c_(x)≦Q(K+M)−1) and LDPC code characteristics can be maintained even if permutation is applied.

FIGS. 33, 34 and 35 illustrate a table showing addresses of parity check matrix according to an embodiment of the present invention. These 3 figures, FIGS. 33, 34 and 35, are parts of one table showing addresses of parity check matrix, but the figure is divided into 3 figures due to lack of space.

The table shown in the FIG. 32 represents a parity check matrix (or matrix H) having a codeword length of 64800 and a code rate of 10/15. The table represents addresses of 1 in the parity check matrix. In this case, the table according to an embodiment of the present invention can be referred to as addresses of a parity check matrix.

In the table of (a), i indicates the blocks generated when the length of the matrix H or codeword by the length of a submatrix. A submatrix according to an embodiment of the present invention is a 360×360 matrix having a codeword length of 64800, and thus the number of blocks can be 180 obtained by dividing 64800 by 360. The each block can be sequentially indicated from 0. Accordingly, i can have a value in the range of 0 to 180. Also i can indicate information bit corresponding to first column in each block.

(b) shows the positions (or addresses) of is (or edges) in the first column in each block.

The matrix H can be represented as H(r,c) using all rows and columns thereof. The following expression 14 is used to derive H(r,c).

$\begin{matrix} {{r = {{\left\lfloor {{x\left( {i,j} \right)}/Q} \right\rfloor \times Q} + {\left( {{x\left( {i,j} \right)} + m} \right){{mod}Q}}}}{{H\left( {r,c} \right)} = \left\{ {{{\begin{matrix} {0,{{{if}\mspace{14mu} r} = {{0\mspace{14mu} {and}\mspace{14mu} c} = 16199}}} \\ {1,\; {else}} \end{matrix}c} = {{i \times Q} + {m\left\lfloor x \right\rfloor}}},{{{the}\mspace{14mu} {largest}\mspace{14mu} {integer}\mspace{14mu} {less}\mspace{14mu} {than}\mspace{14mu} {or}\mspace{14mu} {equal}\mspace{14mu} {to}\mspace{14mu} xj} = 0},\ldots \mspace{14mu},{{length}\mspace{14mu} {of}\mspace{14mu} {x(i)}{m = 0}},\ldots \mspace{14mu},{{Q - {1Q}} = 360}} \right.}} & \left\lbrack {{Expression}\mspace{14mu} 14} \right\rbrack \end{matrix}$

In the expression, X(i,j) represents the j-th value of the i-th line in the table. Specifically, x(0,0)=3454, x(0,1)=3981 and x(1,0)=399, which correspond to the positions of rows having 1 (or addresses of 1) corresponding to i-th line of the matrix H. In this case, maximum values of r and c can be 21599 and 64799, respectively.

The performance of the LDPC code may depend on distribution of degrees of nodes of the parity check matrix, the girth according to the positions of 1s or edges of the parity check matrix, cycle characteristic, connection between check nodes and variable nodes, etc. The matrix H shown optimizes node degree distribution in the case of the codeword of 64800, Q=360 and code rate=10/15 and optimizes the positions of 1s or edges under the condition of optimized degree distribution, Q and code rate.

The matrix H configured according to the table has the above-described QC-IRA LDPC structure. H_(qc) can be obtained using H(r,c) derived using the equation and a base matrix H_(base) can be derived from H_(qc).

In addition, the matrix H according to an embodiment of the present invention may include a matrix H in a different from, which has the same degree distribution as the lengths of x(i) (or degrees of corresponding variable nodes) of the table shown in FIG. 4. Furthermore, when a transmitter performs encoding using the corresponding matrix H, the above-described efficient encoding for QC-IRA LDPC can be employed.

Accordingly, a transmitting side can implement an encoder having high encoding performance, low complexity and high throughput and a receiving side can perform parallel decoding up to 360 level using Q and effectively design a receiver with high throughput using the proposed matrix H.

The following table shows degree distribution.

TABLE 34 Variable node degree 14 9 3 2 (# of variable node)/Q 24 6 90 60

When i is from 0 to 23, the numbers of 1s in the 0-th block to 23rd block are all 14. Accordingly, when the variable node degree is 14, the number of blocks having the same degree is represented as 24. When i corresponds to 120 to 179, the numbers of 1s in the one hundred twentieth to one hundred seventy ninth block are 2. Accordingly, when the variable node degree is 2, the number of blocks having the same degree is 60. As described above, since the parity part of the matrix H includes only submatrices represented as I₀ diagonally distributed in a stepped form, the variable node degree is always 2. Hence, blocks having a variable node degree of 2 can be regarded as blocks corresponding to the parity part. The number of actual variable nodes corresponding to each variable node degree can be obtained by multiplying the number of blocks shown in the table by Q of the submatrix.

FIG. 36 and FIG. 37 illustrate a table showing addresses of parity check matrix according to another embodiment of the present invention. These 2 figures, FIGS. 36 and 37, are parts of one table showing addresses of parity check matrix, but the figure is divided into 2 figures due to lack of space.

The table shown in FIG. 36 and FIG. 37 show the matrix H₂ obtained by modifying the matrix H₁.

In sequential encoding, edges used in a parity processing period are typically represented by an equation and thus the edges can be omitted from the table. That is, 60 blocks having a degree of 2 corresponding to the parity part are not represented in the table.

Since the property of the matrix is maintained even if the matrix is modified, as described above, node degree characteristic, cycle, girth, connection between check nodes and variable nodes, etc. are maintained. Accordingly, the equal encoding performance can be obtained and sequential encoding can be performed using the matrix H₂ according to the table.

FIG. 38 illustrates a method for sequentially encoding the QC-IRA LDPC code according to an embodiment of the present invention.

When the parity check matrix is modified into the matrix H₂ through the above-described permutation process, sequential encoding can be performed through updating of each parity checksum using information bits of a codeword and checksum updating between parity checksums.

As shown in FIG. 38, the codeword can be represented using QK information bits and QM parity checksums. The information bits can be represented as i_(z) according to position and parity checksums can be represented as p_(s).

A parity checksum update process through the information bits can be represented by the following expression 15.

[Expression 15]

p _(w) =p _(w) ⊕i _(z)  (1)

w={v+(z mod Q)M} mod(QM)  (2)

-   -   where z=0, 1, 2, . . . , QK−1

Here, i_(z) represents a z-th information bit and p_(w) denotes a parity checksum that needs to be updated using i_(z). Equation (1) represents that parity checksum p_(w) corresponding to the w-th row is updated through an XOR operation performed on the z-th information and parity checksum p_(w). According to equation (2), the position of w is calculated using the above-described table representing the matrix H₂. Here, v denotes a number corresponding to each row in the table representing the matrix H₂. As described above, a row in the table representing the matrix H₂ corresponds to the position of a block generated when the length of the matrix H or codeword is divided by the submatrix length. Accordingly, the information processing period shown in FIG. 6 is divided by the submatrix length Q and then the numbers of rows corresponding to every Q-th i_(z) are read. Upon completion of checksum update using the information bits of the information processing period, checksum update of the parity processing period can be performed. Checksum update of the parity processing period may be represented by the following expression 16.

p _(s) =p _(s) ⊕p _(s-1) where s=1,2, . . . ,QM−1  [Expression 16]

When S is 0, parity checksum corresponds to parity p₀ and parity values from p₁ to p_(QM-1) can be sequentially derived through XOR operations performed on the parity values and parity values immediately prior thereto.

FIG. 39 illustrates an LDPC decoder according to an embodiment of the present invention.

The LDPC decoder 700 according to an embodiment of the present invention may include a variable node update block 710, a check node update block 720, a barrel shift block 730 and a check sum block 740. Each block will now be described.

The variable node block 710 may update each variable node of the matrix H using an input of the LDPC decoder and a message delivered through edges from the check node block.

The check node block 720 may update a check node of the matrix H using a message transmitted through edges from a variable node. A node update algorithm according to an embodiment of the present invention may include sum product algorithm, belief-propagation algorithm, min-sum algorithm, modified min-sum algorithm, etc. and may be changed according to designer. In addition, since connection between variable nodes and check nodes is represented in the form of a Q×Q circulant identity matrix due to characteristics of QC-IRA LDPC, Q messages between variable nodes and the check node block can be simultaneously processed in parallel. The barrel shift block 730 may control circulant connection.

The check sum block 740 is an optional block which hard-decides a decoding message for each variable node update and performs parity checksum operation to reduce the number of decoding iterations necessary for error correction. In this case, the LDPC decoder 700 according to an embodiment of the present invention can output a final LDPC decoding output through soft decision even if the check sum block 740 hard-decides the decoding message.

FIG. 40, FIG. 41, FIG. 42 and FIG. 43 illustrate one of the embodiments of the H₁ matrix according to a code rate of 7/15.

Above 4 figures are parts of one table which represents H₁ matrix, but the table is divided into 4 figures due to lack of space.

FIG. 44 and FIG. 45 illustrate one of the embodiments of the H₂ matrix according to a code rate of 7/15.

Above 2 figures are parts of one table which represents H₂ matrix, but the table is divided into 2 figures due to lack of space.

FIG. 46 illustrates one of the embodiments of the degree distribution table according to a code rate of 7/15.

One of the embodiments according to a code rate of 8/15 will be described below.

FIG. 47, FIG. 48 and FIG. 49 illustrate one of the embodiments of the H₁ matrix according to a code rate of 8/15.

Above 3 figures are parts of one table which represents H₁ matrix, but the table is divided into 3 figures due to lack of space.

FIG. 50 and FIG. 51 illustrate one of the embodiments of the H₂ matrix according to a code rate of 8/15.

Above 2 figures are parts of one table which represents H₂ matrix, but the table is divided into 2 figures due to lack of space.

FIG. 52 illustrates one of the embodiments of the degree distribution table according to a code rate of 8/15.

One of the embodiments according to a code rate of 11/15 will be described below.

FIG. 53, FIG. 54 and FIG. 55 illustrate one of the embodiments of the H₁ matrix according to a code rate of 11/15.

Above 3 figures are parts of one table which represents H₁ matrix, but the table is divided into 3 figures due to lack of space.

FIG. 56 and FIG. 57 illustrate one of the embodiments of the H₂ matrix according to a code rate of 11/15.

Above 2 figures are parts of one table which represents H₂ matrix, but the table is divided into 2 figures due to lack of space.

FIG. 58 illustrates one of the embodiments of the degree distribution table according to a code rate of 11/15.

One of the embodiments according to a code rate of 12/15 will be described below.

FIG. 59, FIG. 60, FIG. 61 and FIG. 62 illustrate one of the embodiments of the H₁ matrix according to a code rate of 12/15.

Above 4 figures are parts of one table which represents H₁ matrix, but the table is divided into 4 figures due to lack of space.

FIG. 63, FIG. 64, and FIG. 65 illustrate one of the embodiments of the H₂ matrix according to a code rate of 12/15.

Above 3 figures are parts of one table which represents H₂ matrix, but the table is divided into 3 figures due to lack of space.

FIG. 66 illustrates one of the embodiments of the degree distribution table according to a code rate of 12/15.

FIG. 67 and FIG. 68 are a table showing addresses of parity check matrix according to an embodiment of the present invention.

The table shown in FIG. 67 and FIG. 68 represent a parity check matrix (or matrix H) having a codeword length of 16200 and a code rate of 7/15.

Above 2 figures are parts of one table which represents a parity check matrix, but the table is divided into 2 figures due to lack of space.

The table represents addresses of 1 in the parity check matrix. In this case, the table according to an embodiment of the present invention can be referred to as addresses of a parity check matrix.

In the table of (a), i indicates the blocks generated when the length of the matrix H or codeword by the length of a submatrix. The details are as described above.

(b) shows the positions (or addresses) of 1s (or edges) in the first column in each block.

The matrix H can be represented as H(r,c) using all rows and columns thereof

The matrix H shown optimizes node degree distribution in the case of the codeword of 16200, Q=360 and code rate=7/15 and optimizes the positions of 1s or edges under the condition of optimized degree distribution, Q and code rate.

The matrix H configured according to the table has the above-described QC-IRA LDPC structure. The details are as described above. In addition, the matrix H according to an embodiment of the present invention may include a matrix H in a different from, which has the same degree distribution as the lengths of x(i) (or degrees of corresponding variable nodes) of the table shown in FIG. 67 and FIG. 68. The details are as described above.

The following table 35 shows degree distribution.

TABLE 35 Variable node degree 24 4 3 2 (# of variable node)/Q 4 9 8 24

When i is 0 to 3, the numbers of 1s in the 0-th block to 3rd block are all 24. Accordingly, when the variable node degree is 24, the number of blocks having the same degree is represented as 4. When i corresponds to 21 to 44, the numbers of is in the twenty-first block to forty-fourth block are 2. Accordingly, when the variable node degree is 2, the number of blocks having the same degree is 24. The details are as described above.

FIG. 69 is a table showing addresses of parity check matrix according to another embodiment of the present invention.

The table shown in FIG. 69 shows the matrix H₂ obtained by modifying the matrix H₁ having a codeword length of 16200 and a code rate of 7/15.

The details are as described above.

FIG. 70 and FIG. 71 are a table showing addresses of parity check matrix according to another embodiment of the present invention.

The table shown in FIG. 70 and FIG. 71 represent a parity check matrix (or matrix H) having a codeword length of 16200 and a code rate of 9/15.

Above 2 figures are parts of one table which represents a parity check matrix, but the table is divided into 2 figures due to lack of space.

The table represents addresses of 1 in the parity check matrix. In this case, the table according to an embodiment of the present invention can be referred to as addresses of a parity check matrix.

In the table of (a), i indicates the blocks generated when the length of the matrix H or codeword by the length of a submatrix.

The details are as described above.

(b) shows the positions (or addresses) of 1s (or edges) in the first column in each block.

The matrix H can be represented as H(r,c) using all rows and columns thereof.

The performance of the LDPC code may depend on distribution of degrees of nodes of the parity check matrix, the girth according to the positions of 1s or edges of the parity check matrix, cycle characteristic, connection between check nodes and variable nodes, etc. The matrix H shown optimizes node degree distribution in the case of the codeword of 16200, Q=360 and code rate=9/15 and optimizes the positions of 1s or edges under the condition of optimized degree distribution, Q and code rate.

The matrix H configured according to the table has the above-described QC-IRA LDPC structure. The details are as described above. In addition, the matrix H according to an embodiment of the present invention may include a matrix H in a different from, which has the same degree distribution as the lengths of x(i) (or degrees of corresponding variable nodes) of the table shown in FIG. 70 and FIG. 71. The details are as described above. The following table 36 shows degree distribution.

TABLE 36 Variable node degree 16 10 9 3 2 (# of variable 3 2 3 19 18 node)/Q

When i is 0 to 2, the numbers of 1s in the 0-th block to 2nd block are all 16. Accordingly, when the variable node degree is 16, the number of blocks having the same degree is represented as 3. When i corresponds to 27 to 44, the numbers of is in the twenty-seventh block to forty-fourth block are 2. Accordingly, when the variable node degree is 2, the number of blocks having the same degree is 18. The details are as described above.

FIG. 72 is a table showing addresses of parity check matrix according to another embodiment of the present invention.

The table shown in FIG. 72 shows the matrix H₂ obtained by modifying the matrix H₁ having a codeword length of 16200 and a code rate of 9/15.

The details are as described above.

FIG. 73 and FIG. 74 are tables showing addresses of parity check matrix according to another embodiment of the present invention.

The tables shown in the FIG. 73 and FIG. 74 represent a parity check matrix (or matrix H) having a codeword length of 16200 and a code rate of 11/15.

Above 2 figures are parts of one table which represents a parity check matrix, but the table is divided into 2 figures due to lack of space.

The table represents addresses of 1 in the parity check matrix. In this case, the table according to an embodiment of the present invention can be referred to as addresses of a parity check matrix.

In the table of (a), i indicates the blocks generated when the length of the matrix H or codeword by the length of a submatrix.

The details are as described above.

(b) shows the positions (or addresses) of 1s (or edges) in the first column in each block.

The matrix H can be represented as H(r,c) using all rows and columns thereof.

The performance of the LDPC code may depend on distribution of degrees of nodes of the parity check matrix, the girth according to the positions of 1s or edges of the parity check matrix, cycle characteristic, connection between check nodes and variable nodes, etc. The matrix H shown optimizes node degree distribution in the case of the codeword of 16200, Q=360 and code rate=11/15 and optimizes the positions of is or edges under the condition of optimized degree distribution, Q and code rate.

The matrix H configured according to the table has the above-described QC-IRA LDPC structure. The details are as described above.

In addition, the matrix H according to an embodiment of the present invention may include a matrix H in a different from, which has the same degree distribution as the lengths of x(i) (or degrees of corresponding variable nodes) of the table shown in FIG. 73 and FIG. 74. The details are as described above. The following table shows degree distribution.

TABLE 37 Variable node degree 12 3 2 (# of variable node)/Q 7 26 12

When i is 0, 1, 2, 3, 4, 5 and 6, the numbers of is in the 0-th block to 6th block are all 12. Accordingly, when the variable node degree is 12, the number of blocks having the same degree is represented as 7. When i corresponds to 33 to 44, the numbers of 1s in the thirty-third to forty-fourth block are 2. Accordingly, when the variable node degree is 2, the number of blocks having the same degree is 12. The details are as described above.

FIG. 75 is a table showing addresses of parity check matrix according to another embodiment of the present invention.

The table shown in FIG. 75 shows the matrix H₂ obtained by modifying the matrix H₁ having a codeword length of 16200 and a code rate of 11/15.

The details are as described above.

FIG. 76 and FIG. 77 are a table showing addresses of parity check matrix according to another embodiment of the present invention.

The table shown in FIG. 76 and FIG. 77 represent a parity check matrix (or matrix H) having a codeword length of 16200 and a code rate of 5/15.

Above 2 figures are parts of one table which represents a parity check matrix, but the table is divided into 2 figures due to lack of space.

The table represents addresses of 1 in the parity check matrix. In this case, the table according to an embodiment of the present invention can be referred to as addresses of a parity check matrix.

In the table of (a), i indicates the blocks generated when the length of the matrix H or codeword by the length of a submatrix. The details are as described above.

(b) shows the positions (or addresses) of 1s (or edges) in the first column in each block.

The matrix H can be represented as H(r,c) using all rows and columns thereof

The matrix H shown optimizes node degree distribution in the case of the codeword of 16200, Q=360 and code rate=5/15 and optimizes the positions of 1s or edges under the condition of optimized degree distribution, Q and code rate.

The matrix H configured according to the table has the above-described QC-IRA LDPC structure. The details are as described above. In addition, the matrix H according to an embodiment of the present invention may include a matrix H in a different from, which has the same degree distribution as the lengths of x(i) (or degrees of corresponding variable nodes) of the table shown in FIG. 76 and FIG. 77. The details are as described above.

The following table 38 shows degree distribution.

TABLE 38 Variable node degree 30 4 3 2 # of variable node)/Q 3 8 4 30

When i is 0 to 3, the numbers of 1s in the 0-th block to 2-nd block are all 30. Accordingly, when the variable node degree is 30, the number of blocks having the same degree is represented as 3. When i corresponds to 15 to 44, the numbers of is in the twenty-first block to forty-fourth block are 2. Accordingly, when the variable node degree is 2, the number of blocks having the same degree is 30. The details are as described above.

FIG. 78 is a table showing addresses of parity check matrix according to another embodiment of the present invention.

The table shown in FIG. 78 shows the matrix H₂ obtained by modifying the matrix H₁ having a codeword length of 16200 and a code rate of 5/15.

The details are as described above.

FIG. 79 is a table showing addresses of parity check matrix according to another embodiment of the present invention.

The table shown in FIG. 79 represents a parity check matrix (or matrix H) having a codeword length of 16200 and a code rate of 10/15. The table represents addresses of 1 in the parity check matrix. In this case, the table according to an embodiment of the present invention can be referred to as addresses of a parity check matrix.

In the table of (a), i indicates the blocks generated when the length of the matrix H or codeword by the length of a submatrix. The details are as described above.

(b) shows the positions (or addresses) of is (or edges) in the first column in each block.

The matrix H can be represented as H(r,c) using all rows and columns thereof

The matrix H shown optimizes node degree distribution in the case of the codeword of 16200, Q=360 and code rate=10/15 and optimizes the positions of is or edges under the condition of optimized degree distribution, Q and code rate.

The matrix H configured according to the table has the above-described QC-IRA LDPC structure. The details are as described above. In addition, the matrix H according to an embodiment of the present invention may include a matrix H in a different from, which has the same degree distribution as the lengths of x(i) (or degrees of corresponding variable nodes) of the table shown in FIG. 79. The details are as described above.

The following table 39 shows degree distribution.

TABLE 39 Variable node degree 18 4 3 2 (# of variable 5 8 17 15 node)/Q

When i is 0 to 5, the numbers of is in the 0-th block to 4-th block are all 18. Accordingly, when the variable node degree is 18, the number of blocks having the same degree is represented as 5. When i corresponds to 30 to 44, the numbers of is in the twenty-first block to forty-fourth block are 2. Accordingly, when the variable node degree is 2, the number of blocks having the same degree is 15. The details are as described above.

FIG. 80 is a table showing addresses of parity check matrix according to another embodiment of the present invention.

The table shown in FIG. 80 shows the matrix H₂ obtained by modifying the matrix H₁ having a codeword length of 16200 and a code rate of 10/15.

The details are as described above.

FIG. 81 is a table showing addresses of parity check matrix according to another embodiment of the present invention.

The table shown in FIG. 81 represents a parity check matrix (or matrix H) having a codeword length of 16200 and a code rate of 12/15. The table represents addresses of 1 in the parity check matrix. In this case, the table according to an embodiment of the present invention can be referred to as addresses of a parity check matrix.

In the table of (a), i indicates the blocks generated when the length of the matrix H or codeword by the length of a submatrix. The details are as described above.

(b) shows the positions (or addresses) of is (or edges) in the first column in each block.

The matrix H can be represented as H(r,c) using all rows and columns thereof

The matrix H shown optimizes node degree distribution in the case of the codeword of 16200, Q=360 and code rate=12/15 and optimizes the positions of is or edges under the condition of optimized degree distribution, Q and code rate.

The matrix H configured according to the table has the above-described QC-IRA LDPC structure. The details are as described above. In addition, the matrix H according to an embodiment of the present invention may include a matrix H in a different from, which has the same degree distribution as the lengths of x(i) (or degrees of corresponding variable nodes) of the table shown in FIG. 81. The details are as described above.

The following table 40 shows degree distribution.

TABLE 40 Variable node degree 15 4 3 2 (# of variable node)/Q 6 7 23 9

When i is 0 to 5, the numbers of is in the 0-th block to 5-th block are all 15. Accordingly, when the variable node degree is 15, the number of blocks having the same degree is represented as 6. When i corresponds to 36 to 44, the numbers of is in the twenty-first block to forty-fourth block are 2. Accordingly, when the variable node degree is 2, the number of blocks having the same degree is 9. The details are as described above.

FIG. 82 is a table showing addresses of parity check matrix according to another embodiment of the present invention.

The table shown in FIG. 82 shows the matrix H₂ obtained by modifying the matrix H₁ having a codeword length of 16200 and a code rate of 12/15.

The details are as described above.

FIG. 83 illustrates a constellation mapper according to one embodiment of the present invention.

The constellation mapper according to one embodiment of the present invention performs the same operation as the constellation mapper of the BICM block described above.

The data received from the input formatting block described above may be transformed into a bit stream through FEC encoding. In the bit stream, multiple bits constitute a cell, and the cells may be mapped to one of the constellations in the complex plane by the constellation mapper. Herein, for N bits to be transmitted in one cell, 2̂N constellation points may be needed.

Herein, a constellation point may represent one constellation. The constellation point may be referred to as a constellation. 64-QAM, which is a set of constellations, may be called a constellation set, a constellation, and the like.

A constellation may be created using various methods. Depending on the method used to arrange constellation points in a constellation, the probability of errors occurring when the receiver decodes the constellation into a bit stream may vary.

Types of constellations that the constellation mapper uses are as shown in FIGS. 83 (a), 83 (b) and 83 (c). The constellations shown in the figures are exemplary constellations of the respective types. Constellations of FIGS. 83 (a) and 83 (b) are all square QAMs. In the case of FIG. 83 (a), distances between constellation points are non-uniform. In the case of FIG. 83 (b), distances between constellation points are uniform. The constellation of FIG. 83 (a) may correspond to a non-uniform QAM, and the constellation of FIG. 83 (b) may correspond to a normal QAM. The constellation of FIG. 83 (b) may be a special case of FIG. 83 (a).

The present invention proposes constellations as shown in FIG. 83 (c) and a method for finding such constellations. According to the proposed method of the present invention, a lower probability of error, i.e., a higher channel capacity, may be obtained at a given signal-to-noise ratio (SNR) than when the conventional method is used.

FIG. 84 describes a process of mapping IQ-balanced/IQ-symmetric non-uniform constellations according to one embodiment of the present invention.

As another constellation creation method for obtaining the optimum BICM capacities, IQ-balanced/IQ-symmetric non-uniform constellation mapping is proposed.

To find constellation points that maximize the BICM capacity, some assumptions and restraints are needed. Hereinafter, some restraints will be described.

Restraint 1. All constellation points are generated with the same probability. The probabilities of the constellation points may be equal to each other.

Restraint 2. Constellation points do not have a bias. That is, the average of all the constellation points may be 0. In addition, when Restraint 1 is applied, the total sum of the constellation points may be 0.

Restraint 3. The average power of the constellations is a constant. That is, the average power may be invariably set to a constant P.

Restraint 4-1. To implement IQ-balanced mapping, the BICM capacity on the I-axis needs to be equal to the BICM capacity on the Q-axis. For example, if a constellation rotated by a multiple of 90 degrees such as 90, 180 and 270 degrees coincides with the original constellation, this constellation may be viewed as being IQ-balanced. That is, if a constellation point rotated by a multiple of 90 degrees overlaps one of constellation points from an original constellation set, the constellation may be viewed as being IQ-balanced. Hereinafter, a case in which a constellation rotated by a multiple of 90 degrees coincides with the original constellation will be considered as an IQ-balanced mapping scheme.

Restraint 4-2. To implement IQ-symmetric mapping, the BICM capacity in the I-axis should not be equal to the BICM capacity in the Q-axis. To implement IQ-symmetric mapping, constellations should be symmetric with respect to I-axis and Q-axis. For example, when s_(i) is a constellation point of a constellation, conj(s_(i)), −conj(s_(i)), and −s_(i) may also need to be constellation points of the constellation.

Restraint 4-1 and Restraint 4-2 may not be simultaneously met. To implement IQ-balanced mapping, Restraint 4-1 may need to be met. To implement IQ-symmetric mapping, Restraint 4-2 may need to be met. To implement both IQ-balanced mapping and IQ-symmetric mapping, these two restraints need to be met.

Hereinafter, a description will be given of a method of creating non-uniformly distributed constellation points according to IQ-balanced/IQ-symmetric non-uniform constellation mapping.

According to one embodiment, constellation points may be moved in the two-dimensional complex plane to find a constellation providing the optimum capacity. By moving the constellation points, a constellation providing the optimum BICM capacity may be found.

However, moving only one constellation point may not satisfy Restraint 1 and/or Restraint 2 described above. Accordingly, to satisfy the restraints by moving this constellation point, another constellation point may also need to be moved. In this embodiment, the i-th and j-th constellation points are moved.

It is assumed that the constellation prior to moving the constellation points is subject to IQ-balanced mapping. To maintain IQ-balancing, not only the i-th and j-th constellation point pairs but also three other constellation point pairs corresponding thereto may need to be moved. The three corresponding constellation point pairs may refer to constellation points obtained by rotating the i-th and j-th constellation points by 90, 180 and 270 degrees. That is, 8 constellation points may need to be moved together.

The i-th and j-th constellation points to be moved may be defined as s_(0,i) and s_(0,j), and the corresponding constellation points to be moved may be defined as s_(k,i) and s_(k,j). Herein, k=1, 2, 3, and the pairs may respectively represent rotation of s_(0,i) and s_(0,j) by 90, 180 and 270 degrees.

For constellation point s_(0,i), s_(0,j), a and b may be defined as Equations (1) and (2) shown in FIG. 84. Herein, Equations (1) and (2) may respectively mean that Restraint 1 and Restraint 2 are met while the constellations are moved. That is, a and b may be constants. In addition, when s_(0,i) and a are expressed as Equation (3) shown in FIG. 84, Equation (4) may be obtained. Then, Equation (5) may be derived from these equations. It can be seen from Equation (5) that movement of two constellation points can be controlled using one variable θ. Since the other constellation point pairs can be described with s_(0,i), and s_(0,j) as in Equation (6), all the constellation points may be controlled by one variable.

Since movement is implemented with IQ-balancing maintained, the characteristic of IQ-balancing may be maintained even after movement. In this way, a constellation having the optimum capacity may be found with IQ-balancing maintained. When movement is implemented to find the constellation, θ may be split into several parts to calculate the BICM capacity for each split θ to find θ that maximizes the BICM capacity. Using this process, all four sets of s_(i) and s_(j) may be updated.

Hereinafter, specific steps of constructing a constellation having non-uniformly distributed constellation points will be described. Each step may be omitted or replaced with another step, or the sequential order of these steps may change. These steps are intended to describe the spirit of the present invention, not to limit the present invention.

First, an initial constellation may be configured. This initial constellation may be an IQ-balanced or IQ-symmetric constellation. For example, the initial constellation may be a uniform QAM, a non-uniform QAM, or a non-uniform constellation (NUC). Herein, the NUC may be the NUC described above.

Two constellation points (s_(0,i), s_(0,j)) of the initial constellation may be randomly and uniformly selected. The two constellation points should be different from each other. The constellation points may be selected in the first quadrant.

Once the two constellation points are selected, the other constellation points in the second, third and fourth quadrants may also be naturally selected. Accordingly, all eight constellation points may be selected. If the initial constellation is an IQ-balanced, constellation, each of the selected constellations may be represented as e^(jkπ/2). s_(0,i). Here, k=0, 1, 2, 3, and i may be replaced with j. If the initial constellation is an IQ-symmetric constellation, each of the selected constellations may be represented as conj(s_(0,i)), −conj (s_(0,i)) and −s_(0,i). i may be replaced with j. As described above, if the initial constellation is an IQ-balanced or IQ-symmetric constellation, the average of the constellation may be 0.

Thereafter, |s_(0,i)|²+|s_(0,j)|² and the BICM capacity may be calculated. The constellation points may be respectively moved such that the BICM capacity is maximized There may be two methods for finding the optimum constellation position.

One method is to utilize Δ_(i). s_(0,i) may be moved by ±Δ_(i) vertically or horizontally. Accordingly, s_(0,j) may also need to be moved by ±Δ_(j) vertically or horizontally. Herein, Δ_(j) and Δ_(i) may be determined using |s_(0,i)|²+|s_(0,j)|². Accordingly, there may be four cases of movement such as (+Δ_(i), +Δ_(j)), (+Δ_(i), −Δ_(j)), (−Δ_(i), +Δ_(j)), (−Δ_(i), −Δ_(j)). This method may be used for IQ-symmetric non-uniform constellation mapping.

The other method is to utilize θ. As described, movement of constellations may be controlled by θ. Accordingly, by changing θ by a properly small angle each time, a constellation having the optimum capacity may be found. According to an embodiment, the properly small angle may be 1 degree in an embodiment. In addition, the angle may range from 0 degree to 360 degrees. In addition, this range of angle may cover s_(0,i)-a/2 and s_(0,j)-a/2. This is intended to set the optimum capacity within the search range. That is, this is intended to prevent the capacity from being reduced in the search process. This method may be used for IQ-balanced non-uniform constellation mapping.

According to the two methods described above, the BICM capacity may be calculated at constellation positions to which the constellations are respectively moved. If the BICM capacity for the moved constellation is greater than the BICM capacity calculated at first, s_(0,i) and s_(0,j) may be updated with this constellation.

Thereafter, constellations may continue to be searched by reducing Δ_(i) and θ. When these two parameters are sufficiently reduced, two other constellation points of the initial constellation may be selected anew. Then, the optimum position may be found for the newly selected constellation points through the process described above.

When all the BICM capacities are saturated, the algorithm may be stopped and the final constellation set may be obtained. Herein, saturation of the capacities may refer to a case in which great increase in the BICM capacity does not occur in the above algorithm. Saturation of the BICM capacity may be checked every time s_(i) and s_(j) change, or may be checked when all M constellation points are checked.

According to one embodiment, a constellation exhibiting the best performance may be selected after the algorithm is executed for all the seed constellations described above. The seed constellations, namely the initial constellations, may include a uniform QAM, a non-uniform QAM, and a non-uniform constellation (NUC). For example, NUC-64 at the SNR of 10 dB may be obtained by executing the algorithm for the QAM, NUQ and NUC. According to one embodiment, a constellation obtained through the aforementioned algorithm at 9.5 dB or 10.5 dB may be taken as a seed constellation for the algorithm to be executed.

FIG. 85 shows constellations of 64 NUCs at the SNR of 18 dB using the method of IQ-balanced non-uniform constellation mapping according to one embodiment of the present invention.

In this embodiment, uniform-64-QAM having the average power of 1 is taken as a seed constellation, and a desired constellation is found through θ. In addition, the angular increment is set to 1 degree, saturation of the capacity is checked after all M constellation points are updated with s_(i) and s_(j) once. Saturation checking may be performed by checking whether the BICM capacity increases by 1.0e-5 or more. This constellation may remain in the IQ-balanced state and satisfy the n*pi/2-symmetric condition.

To measure the performance of the determined constellation, BICM capacities of the determined constellation may be calculated and compared. For the BICM capacities, Additive White Gaussian Noise (AWGN) and Individually Identical Distributed (IID) input may be assumed. AWGN may represent a basic noise model that is basically used. IID may represent that inputs are independently and equally/uniformly distributed.

$\begin{matrix} {{{BICM}\mspace{14mu} {{cap}.}} = {\sum\limits_{i}\; \left( {{\int_{Y}\ {{p\left( {{b_{i} = 0},y} \right)}\log_{2}\frac{p\left( {{b_{i} = 0},y} \right)}{{p\left( {b_{i} = 0} \right)}{p(y)}}{dy}}} + {\int_{Y}\ {{p\left( {{b_{i} = 1},y} \right)}\log_{2}\frac{p\left( {{b_{i} = 1},y} \right)}{{p\left( {b_{i} = 1} \right)}{p(y)}}{dy}}}} \right)}} & \left\lbrack {{Expression}\mspace{14mu} 17} \right\rbrack \\ \begin{matrix} {{p\left( {{b_{i} = j},y} \right)} = {{p\left( {{yb_{i}} = j} \right)} \cdot {p\left( {b_{i} = j} \right)}}} \\ {= {\sum\limits_{M_{i}}\; {{p\left( {{yx} = M_{j}} \right)} \cdot \frac{1}{M}}}} \\ {= {\sum\limits_{M_{i}}\; {\frac{1}{\pi \; \sigma^{2}}{e^{\frac{- {{y - M_{j}}}^{2}}{\sigma^{2}}} \cdot \frac{1}{M}}}}} \end{matrix} & \left\lbrack {{Expression}\mspace{14mu} 18} \right\rbrack \\ \begin{matrix} {\frac{p\left( {{b_{i} = j},y} \right)}{{p\left( {b_{i} = j} \right)}{p(y)}} = \frac{p\left( {{yb_{i}} = j} \right)}{p(y)}} \\ {= \frac{p\left( {{yb_{i}} = j} \right)}{\sum\limits_{j}\; {p\left( {{b_{i} = j},y} \right)}}} \\ {= \frac{\sum\limits_{M_{i}}\; {\frac{1}{{\pi\sigma}^{2}}{e^{\frac{- {{y - {Mj}}}^{2}}{\sigma^{2}}} \cdot \frac{2}{M}}}}{\sum\limits_{j}\; {\sum\limits_{M_{i}}\; {\frac{1}{{\pi\sigma}^{2}}{e^{\frac{- {{y - {Mj}}}^{2}}{\sigma^{2}}} \cdot \frac{1}{M}}}}}} \end{matrix} & \left\lbrack {{Expression}\mspace{14mu} 19} \right\rbrack \end{matrix}$

The BICM capacity may be calculated using the expression above. Using this expression, values of r and θ that maximize the BICM capacity may be found. Since the AWGN and IID inputs are assumed, it may be expected that y=x+n. Herein, n may denote AWGN. It may be assumed that p(b_(i)=0)=p(b_(i)=1)=1/2. That is, when x is a constellation, and M is a constellation size, it may be possible that p(x=M_(j))=1/M. Herein, M_(j) may be a constellation when b_(i)=j. As shown in FIG. 34, the BICM capacity function may be represented as an integral of a Gaussian function.

The receiver may demap the bits from the constellation. This may be a process reverse to the process of mapping the bits to the constellations described above. LLR may be estimated through demapping. The estimated LLR may be used in the form of a soft input in FEC decoding. In demapping, the process of estimating LLR may be expressed in the following expression.

$\begin{matrix} {{\Pr \left( {b_{i} = j} \right)} = {{\sum\limits_{s_{i} \in \Lambda_{j}}\; {{\Pr \left( {s = s_{k}} \right)}\frac{\exp \left( \frac{{{r - {Hs}_{k}}}^{2}}{\sigma^{2}} \right)}{\sum\limits_{m = 0}^{M - 1}\; {\exp \left( \frac{{{r - {Hs}_{m}}}^{2}}{\sigma^{2}} \right)}}}} = {\sum\limits_{s_{k} \in \Lambda_{j}}\; {{{CPr}\left( {s = s_{k}} \right)}{\exp \left( \frac{{{r - {Hs}_{k}}}^{2}}{\sigma^{2}} \right)}}}}} & \left\lbrack {{Expression}\mspace{14mu} 20} \right\rbrack \\ {{{LLR}\left( b_{i} \right)} = {{\log \left( \frac{\Pr \left( {b_{i} = 1} \right)}{\Pr \left( {b_{i} = 0} \right)} \right)} = {{\log \left( \frac{\sum\limits_{s_{k} \in \Lambda_{l}}\; {{\Pr \left( {s = s_{k}} \right)}{\exp \left( \frac{- {{r - {Hs}_{k}}}^{2}}{\sigma^{2}} \right)}}}{\sum\limits_{s_{k} \in \Lambda_{0}}\; {{\Pr \left( {s = s_{k}} \right)}{\exp \left( \frac{- {{r - {Hs}_{k}}}^{2}}{\sigma^{2}} \right)}}} \right)} = {\log \left( \frac{\sum\limits_{s_{k} \in \Lambda_{l}}\; {\exp \left( \frac{- {{r - {Hs}_{k}}}^{2}}{\sigma^{2}} \right)}}{\sum\limits_{s_{k} \in \Lambda_{0}}\; {\exp \left( \frac{- {{r - {Hs}_{k}}}^{2}}{\sigma^{2}} \right)}} \right)}}}} & \left\lbrack {{Expression}\mspace{14mu} 21} \right\rbrack \\ {{{{LLR}\left( b_{i} \right)} \cong {\frac{{{r - {Hs}_{0}}}^{2}}{\sigma^{2}} - \frac{{{r - {Hs}_{1}}}^{2}}{\sigma^{2}}}} = \frac{{H}^{2}\left( {{2\; {Re}\left\{ {t^{*}s_{0}} \right\}} + {s_{0}}^{2} - {s_{1}}^{2}} \right)}{\sigma^{2}}} & \left\lbrack {{Expression}\mspace{14mu} 22} \right\rbrack \end{matrix}$

Herein, C may be a constant, σ² may be a complex noise power, Λ_(j) may denote a set of constellation points where the i-th bit is j. Herein, j may be 0 or 1. In addition, Pr(s=s_(k)) may denote a priori probability. Here, it may be assumed that b_(i) satisfies equi-probability bit by bit. If iterative decoding is used, the assumption that b_(i) is equi-probable may not be valid, and a priori probability may need to change according to external information from FEC. In addition, when max-log LLR is assumed, it may be possible that t=r/H. s1 may be a constellation that is closest to t for which the i-th bit is 1, and s0 may be a constellation that is closest to t for which the i-th bit is 0.

FIG. 86 shows a constellation of 64 NUCs for 10/15 code rate based on the IQ-balanced non-uniform constellation mapping method according to one embodiment of the present invention.

Coordinates of the constellation points are listed on the right side of the constellation. The respective coordinates represent the coordinates of the constellation points to which bit values are respectively allocated. Each coordinate may be expressed in x and y values for each bit value.

FIG. 87 shows a constellation of 64 NUCs for 10/15 code rate based on the IQ-symmetric non-uniform constellation mapping method according to one embodiment of the present invention.

Coordinates of the constellation points are listed on the right side of the constellation. The respective coordinates represent the coordinates of constellation points to which bit values are respectively allocated. Each coordinate may be expressed in x and y values for each bit value.

FIG. 88 shows a constellation of 256 NUCs for 10/15 code rate based on the IQ-balanced non-uniform constellation mapping method according to one embodiment of the present invention.

FIG. 89 shows a part of the coordinates of the constellations of 256 NUCs for 10/15 code rate based on the IQ-balanced non-uniform constellation mapping method according to one embodiment of the present invention.

FIG. 90 shows another part of the coordinates of the constellations of 256 NUCs for 10/15 code rate based on the IQ-balanced non-uniform constellation mapping method according to one embodiment of the present invention.

FIG. 91 shows the other part of the coordinates of the constellations of 256 NUCs for 10/15 code rate based on the IQ-balanced non-uniform constellation mapping method according to one embodiment of the present invention.

The coordinates represent the coordinates of the constellation points to which bit values are respectively allocated. Each coordinate may be expressed in x and y values for each bit value.

These three coordinate tables, which are separately shown in FIGS. 89 to 82 due to space constraints, constitute one table.

FIG. 92 shows a constellation of 256 NUCs for 10/15 code rate based on the IQ-symmetric non-uniform constellation mapping method according to one embodiment of the present invention.

FIG. 93 shows a part of the coordinates of the constellations of 256 NUCs for 10/15 code rate based on the IQ-symmetric non-uniform constellation mapping method according to one embodiment of the present invention.

FIG. 94 shows another part of the coordinates of the constellations of 256 NUCs for 10/15 code rate based on the IQ-symmetric non-uniform constellation mapping method according to one embodiment of the present invention.

FIG. 95 shows the other part of the coordinates of the constellations of 256 NUCs for 10/15 code rate based on the IQ-symmetric non-uniform constellation mapping method according to one embodiment of the present invention.

The coordinates represent the coordinates of the constellation points to which bit values are respectively allocated. Each coordinate may be expressed in x and y values for each bit value.

These three coordinate tables, which are separately shown in FIGS. 91 and 93 due to space constraints, constitute one table.

Hereinafter, the bit interleaver 5020 according to the present embodiment will be described.

The bit interleaving according to an embodiment of the present invention is designed to achieve optimized performance through combination of the LDPC codes and modulation scheme while providing an efficiently implementable structure.

The bit interleaver 5020 according to the present embodiment may be positioned between the data FEC encoder 5010 and the constellation mapper 5030 as described above to link LDPC-encoded output bits to bit positions having different reliabilities of the constellation mapper 5030 based on LDPC decoding.

As described above with reference to FIG. 23, the bit interleaver 5020 according to the present embodiment may interleave an input bit using parity interleaving, QCB (Quasi-Cyclic Block) interleaving (or block interleaving) and inner group interleaving (or inner-group interleaving).

In addition, the bit interleaver 5020 according to the present embodiment may apply a bit interleaving scheme for acquisition of distribution which is optimal in terms of SNR threshold obtained using density evolution, independently or in addition to the above described bit interleaving scheme.

As described in the foregoing, the bit interleaver 5020 according to the present embodiment is designed to be optimized for the LDPC codes and modulation scheme. Therefore, the present invention proposes bit interleaving parameters for a case in which an LDPC codeword length is 64K or 16K and a combination of a modulation scheme, QPSK, NUC-16, NUC-64, NUC-256, and NUC1K.

Hereinafter, parity interleaving, QCB interleaving and inner group interleaving of the bit interleaver 5020 described with reference to FIG. 23 will be described.

FIG. 96 is a block diagram illustrating a bit interleaver according to an embodiment of the present invention.

The bit interleaver illustrated in FIG. 96 is an example of the above-described bit interleaver 5020, and the bit interleaver according to the present embodiment may include a parity interleaving block 89000, a QCB interleaving block 89100 and an inner group interleaving block 89200. Hereinafter, each block will be described.

The parity interleaving block 89000 may perform interleaving such that bits corresponding to a parity part (parity bits of the FEC block) among LDPC-encoded bits may take a quasi-cyclic (QC) form. In other words, the parity interleaving block 89000 may interleave parity bits in the QC form after parity interleaving, and configure and output QC blocks by collecting bits corresponding to an LDPC QC size. An output of the parity interleaving block 89000 is the same as that described with reference to FIG. 23. In addition, LDPC-encoded bits may be encoded according to the LDCP encoding scheme described with reference to FIGS. 30 to 82.

In the parity interleaving block 89000, parity bits of the FECBLOCK are interleaved by the following expression.

u _(i) =b _(i) for 0≦i<K _(ldpc)

u _(K) _(ldpc) +360×t+s=b _(K) _(ldpc) _(+Q) _(ldpc) ×s+t for

0≦s<360,0≦t<Q _(ldpc)  [Expression 23]

Klpc is a length (or size) of BCH-encoded BBF as illustrated in FIG. 22 and Q_(ldpc) is a code rate dependent constant specified in table 30 and table 31. Note that in the case of QPSK, parity interleaved bits can be directly mapped to the constellation without further interleaving. This can be changed by designer's intention.

The QCB interleaving block 89100 may perform QCB interleaving according to the scheme described with reference to FIG. 23. In other words, as described with reference to FIG. 23, when a plurality of QC blocks output from the parity interleaving block 89000 are input, the QCB interleaving block 89100 may interleave the QC blocks. This is referred to as a QCB interleaving pattern (or interleaving pattern), and the number of QC blocks corresponding to an inner group of inner group interleaving to be performed later is determined based on the QCB interleaving pattern. The QCB interleaving pattern is unique to each combination of modulation type and LDPC code rate.

According to an embodiment of the present invention, an LDPC block having a length of 64800 bits includes 180 (=64800/360) QC blocks, and an LDPC block having a length of 16200 bits includes 45 QC blocks. This can be changed by a designer.

The inner group interleaving block 89200 may perform inner group interleaving according to the scheme described with reference to FIG. 23.

The inner group interleaving block 89200 according to an embodiment of the present invention may perform interleaving according to modulation type and order (η_(mod)) defined in Table 32.

First, the number of QC blocks corresponding to one inner group to be subjected to inner group interleaving needs to be determined to perform inner group interleaving. The bit interleaver 5020 according to the present embodiment may determine the number of QC blocks to configure the inner group using a symmetric reliability of non-uniform QAM (NUQ) and non-uniform constellation (NUC). Specific values are shown in Table 32.

According to an embodiment of the present invention, half of a modulation order is determined to be the number of QC blocks corresponding to the inner group in NUQ or QAM mode, and the number of QC blocks is determined to be the same as the modulation order in NUC. In NUQ, bits corresponding to an I-axis and a Q-axis have the same bit level capacity, and thus only symbol bits corresponding to half of the modulation order have different bit level capacities.

When the number of QC blocks for forming the inner group is determined according to a constellation mode, the inner group interleaving block 89200 according to the present embodiment may perform inner group interleaving thereafter. In this case, QC blocks remaining after the inner group is formed may be present depending on a code rate value. In this case, the inner group interleaving block 89200 according to the present embodiment may successively read bits of four remaining QC blocks not corresponding to the inner groups in an order of bits, and transmit the bits to the constellation mapper 5030. A detailed description thereof will be given below.

The above-described blocks may be omitted or replaced by blocks having similar or identical functions.

Hereinafter, a description will be given of an interleaving process of the inner group interleaving block 89200.

As described above, the inner-group interleaving process is performed with N_(QCB) _(_) _(IG) QC blocks of the QCB interleaving output. Inner-group interleaving is a process of writing and reading (or write and read operation) the bits of the inner-group using 360 columns and N_(QCB) _(_) _(IG) rows. In the write operation, the bits from the QCB interleaving output are written row-wise. The read operation is performed column-wise to read out m bits from each row, where m is equal to 1 for NUC and 2 for NUQ. Therefore, a complexity of the receiver (or a size of buffer in the receiver) can be reduced.

FIG. 97 illustrates an inner group interleaving process for NUQ-256 according to an embodiment of the present invention.

A write operation of inner group interleaving for NUQ-256 is illustrated at the top of FIG. 97, and a read operation of inner group interleaving for NUQ-256 is illustrated at the bottom of FIG. 97.

As illustrated in this figure, the inner group interleaving block 89200 according to the present embodiment may successively write bits corresponding to a QC block to bit 0 to bit 359 of the QC block in an order of bits. In addition, the inner group interleaving block 89200 may cyclically shift and write the bits corresponding to the QC block. This may be changed by a designer.

In addition, as described in the foregoing, in NUQ, the inner group interleaving block 89200 may perform a read operation using 2 bits of each QC block as a unit. As illustrated in the figure, the inner group interleaving block 89200 performs the read operation four times for 2 bits to output 8 bits, and the output 8 bits may be mapped to one symbol by the constellation mapper 5030 that comes after the bit interleaver 5020. A specific operation is the same as that described above, and thus is omitted.

FIG. 98 illustrates a write operation of inner group interleaving for NUC-256 according to an embodiment of the present invention.

Similarly to the above description, in NUC, the inner group interleaving block 89200 may write bits corresponding to each QC block in a row direction, and read the bits in a column direction. In addition, the inner group interleaving block 89200 may cyclically shift and write the bits corresponding to each QC block. This may be changed by a designer.

A specific operation is the same as that described above, and thus is omitted.

FIG. 99 illustrates a read operation of inner group interleaving for NUC-256 according to an embodiment of the present invention.

As described in the foregoing, in NUC, the inner group interleaving block 89200 may perform the read operation using 1 bit of each QC block as a unit. As illustrated in this figure, the inner group interleaving block 89200 may perform the read operation eight times for 1 bit to output 8 bits, and the output 8 bits may be mapped to one symbol by the constellation mapper 5030 that comes after the bit interleaver 5020. A specific operation is the same as that described above, and thus is omitted.

FIG. 100 illustrates a remaining QC block operation according to an embodiment of the present invention.

FIG. 100 illustrates the remaining QC block operation when an LDPC codeword length is 64800 bits and a size of a QC block is 360 bits.

As described in the foregoing, when QC blocks corresponding to an inner group in each modulation are collected, QC blocks not included in any inner group may be present. In the present invention, the QC blocks are referred to as remaining QC blocks. Hereinafter, a description will be given of a table and block diagrams illustrated in the figure.

The table illustrated at the top of the figure indicates the number of remaining QC blocks according to each code rate and modulation. The block diagram illustrated in the middle of the figure indicates the remaining QC blocks during inner interleaving for NUC-256. The block diagram illustrated at the bottom of the figure indicates write and read operations for the remaining QC blocks for NUC-256.

As illustrated in the middle of the figure, in NUC-256, eight QC blocks are collected to form one inner group according to Table 32. When an LDPC codeword length is 64800 bits, 180 (=64800/360) QC blocks in total are generated. Thus, when the 180 QC blocks are divided by 8, 22 inner groups are generated. In this case, while 176 QC blocks may be included in the 22 inner groups, four remaining QC blocks are not included in any inner group and correspond to the remaining QC blocks.

In this case, as illustrated at the bottom of the figure, the inner group interleaving block 89200 according to the present embodiment may successively read bits of four QC blocks in an order of bits, and transmit the read bits to the constellation mapper 5030.

In other words, the remaining 4-QC block bits are mapped to symbols sequentially without performing inner-group interleaving.

FIG. 101 is a block diagram illustrating a demapping & decoding module 9020 according to an embodiment of the present invention.

As described in the foregoing, the demapping & decoding module 9020 may perform a reverse operation of the BICM block 1010 described with reference to FIG. 1.

As illustrated in the figure, the demapping & decoding module 9020 according to the present embodiment may include a cell/time deinterleaver 94000, a demodulator 94100, a bit deinterleaver 94200 and an LDPC/BCH decoder 94300.

First, a symbol passing through a channel is input to the cell/time deinterleaver 94000. The cell/time deinterleaver 94000 may reorder input symbols in a symbol order applied before interleaving. Thereafter, the demodulator 94100 may deduce a log likelihood ratio (LLR) of each bit included in the symbols.

Thereafter, the bit deinterleaver 94200 may perform deinterleaving for reordering input LLR values in an order corresponding to bits to be interleaved. In this case, the bit deinterleaver 94200 may perform inner-group deinterleaving and QCB deinterleaving operations as a reverse operation of the bit interleaver 5020 described above. As illustrated in the figure, the bit deinterleaver 94200 may include an inner group deinterleaving block and a QCB deinterleaving block for performing the above-described deinterleaving.

However, parity deinterleaving corresponding to a reverse operation of parity interleaving may be omitted when decoding is performed based on LDPC bits changed to a QC form in a receiver. In addition, the inner-group deinterleaving and QCB deinterleaving operations described with reference to FIGS. 96 to 99 correspond to reverse operations of read and write operations of interleaving.

The above-described blocks may be omitted or replaced by blocks having similar or identical functions.

FIG. 102 is a block diagram illustrating bit deinterleaving according to another embodiment of the present invention.

FIG. 102 illustrates an example of a case in which, when a QC block interleaving pattern is stored in a ROM of the receiver, the ROM is used, and an LDPC memory and an interleaving memory are shared. This case has an advantage in that a separate memory for bit deinterleaving is not needed.

The bit deinterleaver 94200 and the LDPC/BCH decoder 94300 described with reference to FIG. 101 are illustrated at the top of the figure, and a process of storing the above-described QC block interleaving pattern for bit deinterleaving in the LDPC memory is illustrated at the bottom of the figure.

Specifically, the receiver according to the present embodiment may store an LLR value received through the demodulator 94100 in a register to use the LLR value as an a-priori LLR in an LDPC decoding process. In this case, the number of necessary registers may be determined to be different between NUC and NUQ according to the QC block interleaving pattern. Specifically, in constellation of NUC-256, eight QC blocks are collected to form a NUC-256 symbol, and thus 360×8 registers are needed. However, in NUQ256, inner-group interleaving is performed using QC blocks corresponding to half of a modulation order, and thus 360×4 registers are needed.

Thereafter, the receiver may acquire information about a QC block of LDPC, to which bits correspond, using the interleaving pattern stored in the ROM. Then, the receiver according to the present embodiment may update the LLR value using the information and through CN update. Then, the updated LLR value may be stored in an APP LLR memory again to be used as a priori LLR of a subsequent iteration. A controller illustrated in the figure may direct the above-described process of storing information. When the above-described process is repeated, LDPC decoding may be performed, and deinterleaving may be performed only using the LDPC memory without using a new memory for deinterleaving.

The above-described blocks may be omitted or replaced by blocks having similar or identical functions.

Hereinafter, a description will be given of an operation of the bit interleaver performed when an LDPC codeword length is 64800 bits and a modulation value is NUQ-1024.

FIG. 103 is a table illustrating inner-group interleaving parameters according to another embodiment of the present invention.

The inner-group interleaving according to an embodiment of the present invention is performed according to modulation type and order.

As described in the foregoing, first, the number of QC blocks corresponding to one inner group to be subjected to inner group interleaving needs to be determined to perform inner group interleaving. The bit interleaver 5020 according to the present embodiment may determine the number of QC blocks to configure the inner group using a symmetric reliability of non-uniform QAM (NUQ) and non-uniform constellation (NUC). Information about this is the same as description of Table 32.

In this case, according to an embodiment of the present invention, half of a modulation order may be determined to be the number of QC blocks for the inner group in NUQ or QAM mode, and the same number as the modulation order may be determined to be the number of QC blocks in NUC. In addition, in the present invention, the same number as the modulation order may be determined to be the number of QC blocks subjected to inner group interleaving in NUQ or QAM mode. This may be changed by a designer.

FIG. 103 corresponds to another example of Table 32, and illustrates a table showing a case in which the same number as the modulation order is determined to be the number of QC blocks subjected to inner group interleaving in addition to a case in which a half of the modulation order is determined to be the number of QC blocks for the inner group interleaving in NUQ or QAM mode.

As illustrated in FIG. 103, the number of QC blocks for one inner group may be the same between QAM-16 and NUC-16. In addition, the number of QC blocks for one inner group may be the same between NUC-64 and NUQ-64, between NUC-256 and NUQ-256, and between NUC-1024 and NUQ-1024.

Hereinafter, a description will be given of a bit interleaving process of a case in which the number of QC blocks included in one inner group is the same as the modulation order in NUQ-1024.

FIG. 104 illustrates a write operation of inner group interleaving in NUC-1024 according to an embodiment of the present invention.

As illustrated in this figure, the inner group interleaving block 89200 according to the present embodiment may successively write bits corresponding to QC blocks to bit 0 to bit 359 of the QC blocks in an order of bits. In addition, the inner group interleaving block 89200 may cyclically shift and write the bits corresponding to the QC blocks. This may be changed by a designer.

A specific operation is the same as that described above, and thus is omitted.

FIG. 105 illustrates a read operation of inner group interleaving in NUC-1024 according to an embodiment of the present invention.

As described in the foregoing, in NUC, the inner group interleaving block 89200 may perform the read operation using 1 bit of each QC block as a unit. As illustrated in this figure, the inner group interleaving block 89200 may perform the read operation ten times for 1 bit to output 10 bits, and the output 10 bits may be mapped to one symbol by the constellation mapper 5030 that comes after the bit interleaver 5020. A specific operation is the same as that described above, and thus is omitted.

Hereinafter, a description will be given of an operation of the bit interleaver performed when an LDPC codeword length is 16200 bits and a modulation value is QPSK.

FIG. 106 is a table illustrating inner-group interleaving parameters according to another embodiment of the present invention.

FIG. 106 corresponds to another example of FIG. 103, and the table illustrated in FIG. 106 is the same as the table described with reference to FIG. 103 except that the number of QC blocks subjected to inner group interleaving when a modulation type is QPSK is additionally provided. A specific description is the same as that described above, and thus is omitted.

FIG. 107 illustrates a remaining QC block operation according to another embodiment of the present invention.

FIG. 107 corresponds to another example of the remaining QC blocks illustrated in the middle of FIG. 100, and illustrates remaining QC blocks when an LDPC codeword length is 16200 bits and a size of a QC block is 360 bits.

As described in the foregoing, when QC blocks corresponding to an inner group in each modulation are collected, QC blocks not included in any inner group may be present.

As illustrated in this figure, in NUC-256, eight QC blocks are collected to form one inner group according to the table of FIG. 106. When an LDPC codeword length is 16200 bits, 45(=16200/360) QC blocks are generated in total. Thus, when the number of QC blocks is divided by 8, five inner groups are generated. In this case, while 40 QC blocks in total may be included in five inner groups, five remaining QC blocks are not included in any inner group to become remaining QC blocks.

In this case, as illustrated at the bottom of FIG. 100, the inner group interleaving block 89200 according to the present embodiment may successively read bits of the five QC blocks in an order of bits, and transmit the bits to the constellation mapper 5030. A detailed description thereof is the same as that described above, and thus is omitted.

Hereinafter, a description will be given of bit interleaving to which density evolution is applied.

As described in the foregoing, the bit interleaver 5020 according to the present embodiment may be positioned between the data FEC encoder 5010 and the constellation mapper 5030 to perform a function of spreading output bits of the data FEC encoder 5010 and a function of preventing a multi-edged symbol from being generated through bit twisting (interleaving). The multi-edged symbol (or multi-edge symbol) corresponds to a case in which two or more bits among bits connected to one check node are collected to form one symbol. When the multi-edged symbol is present, LDPC decoding performance on an erasure channel may be degraded. In a conventional terrestrial broadcast system, a column twisting scheme is applied to perform interleaving in order to prevent the multi-edged symbol from being generated at a transmitting end, and a reliability of a modulation bit (MSB or LSB of QAM) to which interleaved bits are connected when the bits are mapped to a QAM symbol by applying demultiplexing after bit interleaving is determined.

Therefore, the conventional terrestrial broadcast system has a problem in that a connection relation between an LDPC bit and a modulation bit connected thereto varies with a demultiplexing pattern (or demuxing pattern), which affects LDPC decoding performance. In addition, the conventional terrestrial broadcast system has a problem in that the demux pattern is fixed to a modulation order value or a value twice the modulation order value, and thus a case thereof is restricted due to a limited modulation order.

In this regard, to overcome the limitations of the conventional terrestrial broadcast system, the present invention proposes bit interleaving having superior performance by obtaining an optimal distribution (connection between blocks in a buffer and a bit) using density evolution.

FIG. 108 is a block diagram illustrating a bit interleaver according to another embodiment of the present invention.

The bit interleaver illustrated in FIG. 108 corresponds to another example of the above-described bit interleaver 5020, and may include a buffer write block 101100 and a cyclic shift operation and buffer read block 102000. As described in the foregoing, the present invention proposes the bit interleaver capable of preventing the multi-edged symbol from being generated using the bit interleaving scheme implemented by an optimal distribution in terms of an SNR threshold obtained using density evolution.

A transmitter or a transmitting device of the present invention may obtain a distribution of an LDPC variable node of each modulation bit after performing bit interleaving by means of density evolution using values of LDPC code information (variable node degree distribution and check node degree distribution) and mutual information (or bitwise capacity) of each modulation bit. In other words, a distribution according to the present embodiment may be defined as a set of variable nodes having the same degree, which is referred to as a degree profile according to an embodiment of the present invention. This may be changed by a designer.

The bit interleaver illustrated in this figure may be designed to maintain a connection corresponding to the degree profile, and thus ensure excellent performance of LDPC decoding at a receiving end.

The bit interleaver illustrated in FIG. 108 corresponds to another example of the above-described bit interleaver 5020, and may include the buffer write block 101100 and the cyclic shift operation and buffer read block 101200. Hereinafter, each block will be described.

The buffer write block 101100 may write bits for interleaving to a buffer according to a result of density evolution. In this case, the buffer write block 101100 may perform writing for each distribution. Thereafter, the cyclic shift operation and buffer read block 101200 may cyclically shift the written bits and then perform reading for each distribution such that the multi-edged symbol is not generated.

The bit interleaver according to the present embodiment uses mutual information of constellation and LDPC information to be subjected to interleaving, and thus may be dependently designed based on each modulation and code.

FIG. 109 corresponds to equations showing a process of obtaining a degree profile through density evolution according to an embodiment of the present invention.

As described in the first block of FIG. 107, the transmitter according to an embodiment of the present invention may calculate bit level capacity (mutual information) for each modulation bit.

Thereafter, as described in the second block of FIG. 107, the transmitter according to an embodiment of the present invention may convert bit level capacity into noise variance.

Thereafter, as described in the second block of FIG. 107, the transmitter according to an embodiment of the present invention may optimize the distribution for each modulation bit by using density evolution.

FIG. 110 illustrates a degree profile according to an embodiment of the present invention.

A matrix L illustrated in this figure corresponds to a degree profile which is a result of density evolution according to the present embodiment. An index j refers to a bit index of modulation, and an index i refers to an index associated with a type of an LDPC variable node degree. A total sum of L_(ij) for respective values of the index i and the index j is 1, which corresponds to a portion of all nodes. In addition, the sum in a column direction corresponds to a ratio of corresponding nodes to the all nodes, and a sum in the row direction corresponds to a ratio of each bit of a symbol to modulation. In 16-QAM, one MSB and one LSB are present in I and Q directions, respectively, and thus a sum in the row direction is 0.25 corresponding to ¼. Thereafter, the actual number of nodes may be obtained by multiplying the matrix L by the entire LDPC length.

FIG. 111 illustrates bit interleaving and bit deinterleaving according to another embodiment of the present invention.

FIG. 111 illustrates bit interleaving and bit deinterleaving when a codeword length is 24K and a modulation type is QAM-16.

Specifically, a process of performing bit interleaving on a buffer expressed as blocks is illustrated on the left side of the figure, and a process of performing bit deinterleaving on a buffer expressed as blocks is illustrated on the right side of the figure.

In addition, bit interleaving illustrated on the left side of the figure corresponds to a case in which only write and read operations are performed except for cyclic-shift for avoiding the above-described multi-edged symbol and a configuration of writing and reading of interleaver/deinterleaver. Bit deinterleaving corresponding thereto is illustrated on the right side of the figure.

Hereinafter, a description will be given by focusing on bit interleaving illustrated on the left side of the figure.

In the present invention, the number of memory buffers for bit interleaving in the row direction is the same as a modulation order, and the number of columns is the same as a value obtained by dividing the entire codeword length by the modulation order. In addition, each row of the buffer illustrated in this figure corresponds to each bit of a modulation symbol, and each row may correspond to an MSB or an LSB. The number of optimal variable node degrees of each modulation bit may be obtained through density evolution described with reference to FIGS. 108 and 109, and each bit may be written in a distribution according to a corresponding degree value. Thus, blocks colored different colors in the buffer correspond to a set, that is, a distribution of variable nodes having different degrees. In addition, the numbers of bits of variable node degrees of first MSBs are 2, 2, and 2, respectively.

The above-described bit interleaver or buffer write block 101100 may write input bits to a buffer for each distribution. Thereafter, the above-described bit interleaver or cyclic shift operation and buffer read block 101200 may read four bits of the buffer in the column direction. In this case, the four bits may be mapped to one symbol in the constellation mapper 5030.

Bit deinterleaving illustrated on the right side of the figure corresponds to a reverse operation of the above-described bit interleaving, and includes write and read operations in which input bits are continuously written to a deinterleaving memory in the column direction. In this case, the read operation is performed for each distribution, and thus a sequence generated before bit interleaving may be restored.

An input sequence of bit interleaving, an output sequence of bit interleaving, an input sequence of bit deinterleaving and an output sequence of bit deinterleaving are illustrated at the bottom of the figure.

As illustrated in the figure, in the input sequence of bit interleaving, bits corresponding to successive figures, that is, bit 1, bit 2, bit 3, . . . may be successively written to each distribution. Thereafter, when bit interleaving is performed, bit 1, bit 3, bit 5 and bit 6 in the column direction are successively output. The input sequence of bit deinterleaving is the same as the output sequence of bit interleaving, and the output sequence of bit deinterleaving is the same as the input sequence of bit interleaving.

FIG. 112 illustrates a bit interleaving pattern according to an embodiment of the present invention.

When one symbol is generated, if bits corresponding to parity parts of an LDPC FEC block are mixed, performance on the erasure channel may be enhanced. Therefore, the present invention proposes bit interleaving (or a bit interleaving pattern) capable of generating a symbol in which parity parts are mixed when the bit interleaving pattern is generated through DPM density evolution.

Specifically, the present invention determines an amount of an optimal parity portion part of each bit of a symbol in density evolution, and proposes an interleaving pattern in which interleaving may be performed such that portions of respective parity parts do not overlap.

Similarly to FIG. 111, FIG. 112 illustrates bit interleaving corresponding to a case in which a codeword length is 24K and a modulation type is QAM-16, and illustrates a case in which an MSB includes two parity parts (low degree nodes) and an LSB includes one parity part through density evolution.

Blocks on the left side of the figure illustrate positions of respective variable nodes set according to a distribution in a buffer. As described above, blocks colored different colors correspond to a set, that is, a distribution of variable nodes having different degrees. Circles in the left blocks indicate parity portion parts in the buffer.

Blocks on the right side of the figure illustrate a case in which parity parts are uniformly dispersed through interleaving. Circles in the right blocks indicate parity portion parts uniformly dispersed in the buffer.

In particular, in the present invention, the bit interleaving pattern may be adjusted such that parity parts are uniformly dispersed through cyclic shifting. In addition, parity parts may be uniformly dispersed in a symbol using a scheme other than a cyclic shifting scheme according to a designer. Thereafter, the bit interleaver according to the present embodiment may perform writing configuration search and/or cyclic shifting for removing the multi-edged symbol. Specific description will be given below.

Hereinafter, a description will be given of a writing configuration according to an embodiment of the present invention.

As described in the foregoing, the bit interleaver according to the present embodiment may write input bits for each distribution. In this case, the bit interleaver may write the input bits according to an MSB or an LSB in a symbol of QAM. This may be referred to as the writing configuration. The bit interleaver according to the present embodiment may select one writing configuration from among a plurality of writing configurations, and write corresponding bits according to the selected writing configuration.

FIG. 113 illustrates a writing configuration according to an embodiment of the present invention.

As described in the foregoing, when the number (distribution) of nodes having the same particular degree for respective rows corresponding to an MSB and an LSB is determined, the bit interleaver according to the present embodiment may write bits to a buffer based on the determination. Therefore, the writing configuration according to the present embodiment may be determined based on the specific number of bits of each of the MSB and the LSB.

A graph at the top of FIG. 113 illustrates a relation between a distribution of variable nodes (Dv) and the variable node. Boxes on the right side of the figure illustrate a writing configuration of 16-QAM and a writing configuration of 64-QAM.

Blocks at the bottom of FIG. 113 are an example illustrating a writing order when LMML is selected as a high degree variable node configuration from among a plurality of writing configurations of 16-QAM.

As a result, the writing configuration indicates a connection between bits and the MSB and the LSB. This specific connection affects performance of LDPC decoding or the receiver, and thus the bit interleaver according to the present embodiment may perform the write operation of interleaving by selecting an appropriate configuration based on a combination of a code rate and each modulation.

Specifically, when MMLL is selected from among writing configurations of 16-QAM illustrated in the figure, bits 1 to 10 may be connected to MSB1, bits 11 to 20 may be connected to MSB2, and bits 21 to 25 and bits 26 to 30 may be connected to LSB1 and LSB2, respectively, in the corresponding distribution.

In addition, when MLML is selected from among writing configurations of 16-QAM illustrated in the figure, bits 1 to 10 may be connected to MSB1, bits 11 to 15 may be connected to LSB1, and bits 16 to 25 and bits 25 to 30 may be connected to MSB2 and LSB4, respectively, in the corresponding distribution.

The writing configuration according to the present embodiment may be differently set between regions (distributions) in which respective variable nodes are included, and set by determining the regions to be M1-1, M1-2, M2-1, M2-2, L1-1, L1-2, L2-1, L2-2, and the like obtained by subdividing four regions of M1M2L1L2. Alternatively, a specific position may be determined for each bit. This may be changed by a designer.

FIG. 114 illustrates a cyclic shift operation according to an embodiment of the present invention.

As described in the foregoing, the multi-edged symbol is a main factor that degrades LDPC decoding performance on the erasure channel, and thus bit interleaving needs to be performed such that a multi-edged symbol is not generated.

Blocks on the left side of the figure illustrate a write operation performed before cyclic shifting, and blocks on the right side of the figure illustrate a read operation performed after cyclic shifting.

Hereinafter, it is presumed that bit 4 and bit 11 corresponding to a circle indicated in the left blocks of the figure are bits which are in the same check node and may generate the multi-edged symbol.

The bit interleaver or the cyclic shift operation and buffer read block 101200 according to the present embodiment may detect whether the multi-edged symbol is generated using a given matrix H and the bit interleaving pattern.

In this case, the bit interleaver or the cyclic shift operation and buffer read block 101200 according to the present embodiment may identify a cyclic shift factor for shifting corresponding bits. Thereafter, the bit interleaver or the cyclic shift operation and buffer read block 101200 according to the present embodiment may reconfigure a buffer memory value by performing row-wise cyclic shifting of the corresponding bits using the selected cyclic shift factor, and then perform the read operation. Through this process, it is possible to prevent the multi-edged symbol from being generated.

The cyclic shift factor is [0 0 1 0] in the present embodiment, and used to avoid generation of the multi-edged symbol of a total of six (=24/4) symbols. As illustrated in the figure, first, second and fourth rows are not shifted and only a third row is shifted in the right direction according to the selected cyclic shift factor. Cyclic shifting may be performed either clockwise or counterclockwise according to intention of a designer.

When the cyclic shift operation is performed in bit interleaving, a cyclic shift operation corresponding to a reverse direction of cyclic shifting of bit interleaving may be performed after the write operation in bit deinterleaving.

Hereinafter, a description will be given of a case in which bit interleaving described with reference to FIGS. 106 to 113 is applied using a QC block as a unit.

FIG. 115 illustrates bit interleaving with a QC level according to an embodiment of the present invention.

As described in the foregoing, LDPC encoding having a QC form may be used to enable the receiver to perform parallel decoding.

In QC LDPC encoding, bits included in one QC form (or QC block) have the same connectivity and are connected to a check node the same number of times (have the same variable node degree). Therefore, the bit interleaver may perform the write operation using a QC block as a unit (or a QC level) instead of using a bit as a unit. It means every bit in same QC can be connected to the same level (for example MSB or LSB).

Blocks of the figure illustrate a buffer subjected to bit interleaving using a QC bock as a unit. A number of each QC block illustrated in the figure indicates a set of bits corresponding to each QC. Specifically, 3QC indicates a set of bits included in a third QC block. When a QC block has a size of 360, 360 bits may be included in the block.

When the bit interleaver according to the present embodiment reads written QC blocks in the column direction, the bit interleaver may output the read QC blocks to the constellation mapper 5030. Bits included in the output QC blocks may generate a final symbol.

In addition, the bit interleaver may be designed using ½ QC block as a unit according to a relation between a total number of QC blocks and a modulation order. This may be changed by a designer.

FIG. 116 illustrates a cyclic shift operation with a QC level and a bit level according to an embodiment of the present invention.

FIG. 116 corresponds to another example of the cyclic shift operation described with reference to FIG. 114.

The cyclic shift operation according to an embodiment of the present invention may also be adapted to QC level bit interleaving. In this case, the cyclic shift factor with QC level cannot avoid whole multi-edged symbol. Therefore the cyclic shift with bit level may be adapted to each QC block.

Blocks at the top of the figure are blocks indicating a buffer subjected to bit interleaving using a QC block as a unit as described above. Circles on the blocks indicate QC blocks in which the multi-edged symbol may be generated. In this case, as illustrated in the figure, cyclic shifting may be performed on a third row using a QC block as a unit (or a QC level).

Blocks in the middle of the figure are blocks indicating a buffer subjected to cyclic shifting performed using a QC block as a unit. A circle on the blocks indicates a case in which a multi-edged symbol is generated even when cyclic shifting is performed using a QC block as a unit. In this case, cyclic shifting is performed using a bit as a unit (or a bit level) within 12 QC blocks.

Blocks at the bottom of the figure are blocks indicating a buffer subjected to cyclic shifting performed using a bit as a unit. A block 12QC′ illustrated in this figure indicates a block 12QC subjected to cyclic shifting using a bit as a unit.

Cyclic shifting using a QC block as a unit and cyclic shifting using a bit as a unit may be performed either clockwise or counterclockwise according to intention of a designer.

FIG. 117 illustrates a cyclic shift operation for evenly dispersed parity parts according to an embodiment of the present invention.

As described above, for erasure channel, the transmitter or the bit interleaver according to an embodiment of the present invention can disperse parity parts evenly and then the transmitter or the bit interleaver according to an embodiment of the present invention may operate writing configuration search to decide the optimized variable node position. However, it can cause the multi edged symbol.

In this case, the transmitter according to an embodiment of the present invention or the bit interleaver according to an embodiment of the present invention can perform simple cyclic shift operation without the parity part. By using this operation, the transmitter or the bit interleaver can avoid the multi-edged symbol.

The bit interleaver according to the present embodiment may uniformly disperse the parity parts described with reference to FIG. 112, determine a position of an optimal variable node through the writing configuration search described with reference to FIG. 113, and then verify whether the multi-edged symbol is generated. In this case, when the multi-edged symbol is generated, parity parts, which are uniformly dispersed when cyclic shifting is applied according to FIG. 114 and the scheme described with reference to FIG. 114, may not be uniformly dispersed. Thus, the present invention proposes a scheme of applying cyclic shifting only to remaining variable nodes except for the parity parts. In this way, it is possible to maintain a distribution of uniformly dispersed parity parts, and prevent the multi-edged symbol from being generated after bit interleaving.

Blocks on the left side of the figure are blocks indicating a buffer corresponding to a case in which cyclic shifting is performed except for the parity parts, and blocks on the right side of the figure are blocks indicating a buffer after cyclic shifting is performed.

Circles on the left blocks of the figure indicate bits resulting in the multi-edged symbol. In this case, the bit interleaver according to the present embodiment may perform cyclic shifting by selecting an appropriate cyclic shift factor. In the left blocks of the figure, the cyclic shift operation is applied to a third row and a fourth row counterclockwise.

In this case, bit 21 and bit 22 included in the third row and the fourth row correspond to parity parts.

Thus, as illustrated in the right blocks of the figure, the bit interleaver according to the present embodiment may not perform the cyclic shift operation on bit 21 and bit 22, and perform the cyclic shift operation only on remaining bits. In this case, bits colored the same color as that of bit 21 and bit 22 correspond to parity parts, and thus the multi-edged symbol is not generated while a distribution of uniformly dispersed parity parts is maintained.

FIG. 118 is a block diagram illustrating the demapping & decoding module 9020 according to another embodiment of the present invention.

FIG. 118 is another example of the demapping & decoding module 9020 described with reference to FIG. 101. As described in the foregoing, the demapping & decoding module 9020 may perform a reverse operation of the BICM block 1010 described with reference to FIG. 1.

As described in this figure, the demapping & decoding module according to another embodiment of the present invention may include a cell/time deinterleaver block, a demodulator, a bit deinterleaver 111000 and an LDPC/BCH decoder. Except for the bit deinterleaver 111000, blocks may perform the same operations as those described with reference to FIG. 101, and thus specific descriptions thereof are omitted.

The bit deinterleaver 111000 is another example of the bit deinterleaver 92200 described with reference to FIG. 101, and may include a buffer write and cyclic reverse shift operation block 111100 and a buffer read using DE result block 111200. In addition, the bit deinterleaver 111000 may perform a reverse operation of bit interleaving described with reference to FIGS. 108 to 117. Specific description thereof is omitted.

FIG. 119 illustrates a DE result and a buffer associated with the DE result according to an embodiment of the present invention.

Specifically, the DE result corresponding to a case in which a modulation type is QAM-256 and a code rate is 3/4 is illustrated at the top of the figure, and the buffer associated with the DE result is illustrated at the bottom of the figure.

The table illustrated at the top of the figure may be referred to as a degree profile. When a modulation type is QAM-256 and a code rate is 3/4, three types of variable nodes, that is, a variable node having a degree of 8, a variable node having a degree of 3, and a node having a degree of 2 or 1 (the node having the degree 1 corresponds to the node having the degree 2) may be present. The table at the top of the figure illustrates a connection between each variable node and each bit (MSB, LSB) of a 256QAM symbol.

As illustrated on the right side at the top of the figure, when writing configurations for the variable node having the degree of 3 and the variable node having the degree of 2 are determined, the bit interleaver according to the present embodiment may write each bit to a buffer based on the writing configurations and the DE result.

A block illustrated at the bottom of the figure indicates a result obtained when bits are written to the buffer. The bit interleaver according to the present embodiment may read the bits written to the buffer in a column direction, and finally output the bits. In this case, the output bits may be successively connected to positions corresponding to M1 M2 A1 A2 A3 A4 L1 L2 QAM bits.

FIG. 120 is a graph illustrating bit error rate (BER) and block error rate (BLER) results of a symbol according to an embodiment of the present invention.

Specifically, FIG. 120 is a graph illustrating a BER and a BLER obtained when a symbol, to which bit interleaving described above is applied, passes through an additive white Gaussian noise (AWGN) channel, is demodulated, and then is LDPC-decoded through deinterleaving.

FIG. 121 is a flowchart illustrating a method for transmitting broadcast signals according to an embodiment of the present invention.

The apparatus for transmitting broadcast signals (or a transmitter) according to an embodiment of the present invention can encode service data (S114000). As described above, service data is transmitted through a data pipe which is a logical channel in the physical layer that carries service data or related metadata, which may carry one or multiple service(s) or service component(s). Data carried on a data pipe can be referred to as the DP data or the service data. The detailed process of step S114000 is as described in FIG. 1 or FIG. 5-6, FIG. 22, FIG. 30-FIG. 82.

The apparatus for transmitting broadcast signals according to an embodiment of the present invention can bit interleaving the encoded service data (S114010). The bit interleaving is designed to achieve optimized performance with combination of the LDPC codes and modulation scheme while providing an efficiently implementable structure. The outputs of the LDPC encoder are bit-interleaved, which consists of parity interleaving followed by Quasi-Cyclic Block (QCB) interleaving and inner-group interleaving. The detailed process of the bit interleaving is as described in FIG. 96-FIG. 120.

The transmitter according to an embodiment of the present invention may map the bit interleaved service data into a plurality of OFDM symbols to build at least one signal frame (S114020). The detailed process of this step is as described in FIG. 7, FIG. 10-21.

Then the transmitter according to an embodiment of the present invention may data in the built at least one signal frame by an OFDM scheme (S114030).

Subsequently, the transmitter according to an embodiment of the present invention may transmit the broadcast signals including the modulated data (S114040). The detailed process of this step is as described in FIG. 1 or 8.

FIG. 122 is a flowchart illustrating a method for receiving broadcast signals according to an embodiment of the present invention.

The flowchart shown in FIG. 122 corresponds to a reverse process of the broadcast signal transmission method according to an embodiment of the present invention, described with reference to FIG. 121.

The apparatus for receiving broadcast signals (or a receiver) according to an embodiment of the present invention can receive broadcast signals (S115000).

The apparatus for receiving broadcast signals according to an embodiment of the present invention can demodulate the received broadcast signals using an OFDM (Orthogonal Frequency Division Multiplexing) scheme (S115010). Details are as described in FIG. 9.

The receiver (or the apparatus for receiving) according to an embodiment of the present invention may parse a signal frame from the demodulated broadcast signals (S115020). Details are as described in FIG. 9.

Subsequently, the receiver according to an embodiment of the present invention may bit deinterleave data in the parsed signal frame (S115030). In this case, the receiver according to an embodiment of the present invention can perform bit deinterleaving as a reverse processing of the bit interleaving. The detailed process of the bit interleaving is as described in FIG. 96-FIG. 121.

The receiver according to an embodiment of the present invention may decode the bit deinterleaved data (S115040). Details are as described in FIG. 1. FIG. 5-6, FIG. 9, FIG. 22 and FIG. 30-FIG. 82.

As described above, service data is transmitted through a data pipe which is a logical channel in the physical layer that carries service data or related metadata, which may carry one or multiple service(s) or service component(s). Data carried on a data pipe can be referred to as the DP data or the service data.

Herein, the bit interleaving pattern table indicating a bit interleaving pattern according to a code word length, modulation type and code rate will be described. As above described, the bit interleaving pattern is unique to each combination of modulation type and LDPC code rate. The bit interleaving pattern can be referred as a QCB interleaving pattern.

Each QCB interleaving pattern table shows a relationship between the QCB interleaving (group wise interleaving) output and the QCB interleaving (group wise interleaving) input according to at least one specific code rate. As above described, the output of the LDPC encoding can be divided into 180 QC blocks and each QC block may be represented as a number of 0-179. Therefore, when the LDPC code word length is 64800, 0-359 bits can correspond to 0^(th) QC block and 360-719 bits can correspond to 1^(st) QC block.

The first column of each table shows the order of QCB interleaving (group wise interleaving) output and each of the other column shows the order of the QCB interleaving (group wise interleaving) input according to a specific code rate.

Table 41 represents a QCB interleaving pattern table when the modulation type is NUC 16 and the code word length is 16K.

TABLE 41 Code rate 6/15 7/15 9/15 11/15 13/15 j-th block π(j)-th π(j)-th π(j)-th π(j)-th π(j)-th of Group- block of block of block of block of block of wise Group- Group- Group- Group- Group- Inter- wise wise wise wise wise leaver Interleaver Interleaver Interleaver Interleaver Interleaver output input input input input input 0 12 19 4 2 12 1 13 3 6 4 7 2 15 32 19 41 20 3 30 38 2 8 43 4 27 16 5 13 29 5 25 17 30 7 13 6 11 29 20 0 32 7 34 33 11 24 30 8 9 14 22 3 25 9 4 10 12 22 0 10 31 6 15 5 17 11 22 2 0 32 18 12 6 20 36 10 9 13 32 15 37 9 1 14 7 40 38 36 41 15 21 39 39 37 42 16 17 12 26 29 6 17 3 22 14 11 33 18 1 23 34 25 28 19 26 34 35 16 14 20 10 31 16 20 16 21 33 13 13 21 11 22 19 44 18 35 39 23 2 43 42 34 40 24 18 36 7 15 15 25 5 24 10 1 4 26 28 37 25 6 23 27 35 42 43 14 5 28 8 0 40 27 2 29 16 9 17 30 24 30 29 4 41 33 22 31 23 21 24 12 38 32 14 5 33 17 10 33 0 35 31 28 8 34 20 26 23 23 19 35 24 41 32 40 34 36 36 7 21 26 26 37 37 28 3 31 36 38 38 11 27 38 37 39 39 25 28 39 27 40 40 8 8 18 21 41 41 18 9 19 31 42 42 1 29 42 3 43 43 30 1 43 35 44 44 27 44 44 44

Table 42 represents a QCB interleaving pattern table when the modulation type is NUC 64 and the code word length is 16K.

TABLE 42 Code rate 6/15 7/15 9/15 π(j)-th π(j)-th π(j)-th j-th block of block of block of block of Group-wise Group-wise Group-wise Group-wise Interleaver Interleaver Interleaver Interleaver output input input input 0 31 2 21 1 12 14 5 2 39 10 43 3 32 0 38 4 30 37 40 5 24 42 1 6 28 38 3 7 15 40 17 8 38 24 11 9 23 29 37 10 27 28 10 11 41 35 41 12 0 18 9 13 6 16 15 14 17 20 25 15 37 27 44 16 42 41 14 17 20 30 27 18 11 15 7 19 4 19 18 20 40 9 20 21 2 43 35 22 3 25 16 23 26 3 0 24 10 6 6 25 7 7 19 26 13 31 8 27 25 32 22 28 1 26 29 29 18 36 28 30 8 17 34 31 5 1 31 32 14 13 33 33 36 5 30 34 35 39 32 35 33 33 42 36 22 4 13 37 9 8 4 38 44 23 24 39 16 22 26 40 34 11 36 41 19 34 2 42 21 44 23 43 29 12 12 44 43 21 39

Table 43 represents a QCB interleaving pattern table when the modulation type is NUC 256 and the code word length is 16K.

TABLE 43 Code rate 6/15 11/15 π(j)-th π(j)-th j-th block of block of block of Group-wise Group-wise Group-wise Interleaver Interleaver Interleaver output input input 0 17 8 1 13 13 2 25 0 3 24 11 4 14 9 5 21 4 6 1 36 7 37 37 8 2 16 9 3 3 10 11 10 11 22 14 12 18 24 13 5 20 14 10 33 15 23 34 16 12 25 17 4 2 18 26 21 19 16 31 20 38 12 21 36 19 22 33 7 23 39 5 24 0 27 25 6 23 26 7 26 27 31 1 28 32 18 29 34 22 30 27 35 31 35 6 32 15 32 33 9 30 34 30 28 35 28 15 36 19 29 37 8 17 38 20 39 39 29 38 40 40 40 41 41 41 42 42 42 43 43 43 44 44 44

Table 44 represents a QCB interleaving pattern table when the modulation type is QPSK and the code word length is 16K.

TABLE 44 Code rate 6/15 7/15 9/15 j-th π(j)-th π(j)-th π(j)-th block block of block of block of of Group- Group- Group- Group- wise wise wise wise Interleaver Interleaver Interleaver Interleaver output input input input 0 7 3 0 1 4 7 1 2 0 1 2 3 5 4 3 4 27 18 4 5 30 21 5 6 25 22 6 7 13 6 7 8 31 9 8 9 9 5 9 10 34 17 10 11 10 14 11 12 17 13 12 13 11 15 13 14 8 10 14 15 12 20 15 16 15 8 16 17 16 19 17 18 18 16 18 19 19 12 19 20 20 0 20 21 21 11 21 22 22 2 22 23 23 23 23 24 1 24 24 25 35 25 25 26 24 26 26 27 29 27 27 28 33 28 28 29 6 29 29 30 26 30 30 31 14 31 31 32 32 32 32 33 28 33 33 34 2 34 34 35 3 35 35 36 36 36 36 37 37 37 37 38 38 38 38 39 39 39 39 40 40 40 40 41 41 41 41 42 42 42 42 43 43 43 43 44 44 44 44

Table 45 represents a QCB interleaving pattern table when the modulation type is NUC16 and the code word length is 64K.

TABLE 45 Code rate 5/15 8/15 9/15 j-th π(j)-th π(j)-th π(j)-th block block of block of block of of Group- Group- Group- Group- wise wise wise wise Interleaver Interleaver Interleaver Interleaver output input input input 1 65 81 89 2 160 170 10 3 140 101 142 4 32 143 19 5 50 77 41 6 162 128 1 7 86 112 146 8 177 155 68 9 57 41 87 10 157 40 9 11 9 54 51 12 134 57 114 13 104 28 92 14 24 179 121 15 7 114 69 16 122 97 107 17 46 13 97 18 17 18 166 19 77 151 162 20 31 91 55 21 92 88 174 22 163 79 126 23 148 92 149 24 133 137 110 25 99 27 128 26 18 122 172 27 0 107 28 28 167 135 111 29 101 82 78 30 110 125 82 31 135 103 120 32 124 74 71 33 71 36 52 34 107 9 5 35 5 93 141 36 123 0 29 37 69 86 30 38 108 63 132 39 141 158 148 40 179 148 72 41 96 25 85 42 113 167 17 43 83 116 160 44 176 70 156 45 52 43 154 46 117 102 131 47 81 106 164 48 125 149 65 49 59 24 76 50 15 169 125 51 137 113 50 52 170 127 16 53 63 34 130 54 112 165 129 55 88 100 143 56 34 136 133 57 61 75 98 58 106 134 0 59 3 156 42 60 42 96 63 61 100 84 83 62 152 178 173 63 87 150 49 64 171 140 74 65 72 20 43 66 161 126 8 67 4 73 147 68 178 68 61 69 64 130 36 70 150 121 167 71 10 48 119 72 128 53 27 73 49 22 86 74 26 129 102 75 75 99 48 76 41 11 115 77 102 33 99 78 28 124 38 79 2 157 163 80 168 161 73 81 93 29 101 82 156 123 4 83 12 160 153 84 38 55 118 85 45 26 90 86 151 168 124 87 142 98 151 88 44 67 66 89 66 15 93 90 25 7 123 91 139 94 157 92 173 144 24 93 51 1 44 94 29 61 168 95 147 65 80 96 175 146 15 97 90 42 39 98 164 172 178 99 80 115 45 100 131 59 21 101 58 76 37 102 114 4 11 103 145 162 136 104 121 39 113 105 70 85 77 106 115 12 122 107 146 72 158 108 120 58 64 109 55 44 81 110 158 132 6 111 8 47 60 112 39 141 54 113 97 35 35 114 159 176 13 115 138 104 57 116 33 139 171 117 47 80 100 118 116 6 117 119 79 95 46 120 174 87 62 121 74 90 33 122 21 173 175 123 6 163 137 124 130 69 59 125 54 32 103 126 109 8 127 127 76 154 70 128 35 145 108 129 98 23 88 130 155 177 179 131 144 111 40 132 36 60 112 133 94 38 104 134 23 171 170 135 78 62 140 136 165 46 67 137 56 21 32 138 154 5 105 139 89 153 159 140 132 49 26 141 67 78 96 142 119 2 169 143 143 109 135 144 40 147 109 145 53 89 47 146 20 166 177 147 136 152 56 148 172 138 116 149 91 31 79 150 27 14 106 151 13 131 150 152 127 50 25 153 73 37 94 154 105 16 134 155 85 117 152 156 30 66 22 157 103 19 84 158 19 10 176 159 84 159 139 160 37 142 20 161 48 105 34 162 153 3 165 163 11 164 138 164 166 51 7 165 60 83 91 166 111 174 12 167 14 108 145 168 169 52 58 169 95 17 95 170 118 64 2 171 1 119 144 172 126 45 53 173 68 133 75 174 22 175 14 175 149 110 155 176 43 56 18 177 62 30 31 178 16 120 3 179 82 118 161

Table 46 represents a QCB interleaving pattern table when the modulation type is NUC 64 and the code word length is 64K.

TABLE 46 Code rate 7/15 9/15 10/15 13/15 j-th block π(j)-th π(j)-th π(j)-th π(j)-th of block of block of block of block of Group-wise Group-wise Group-wise Group-wise Group-wise Interleaver Interleaver Interleaver Interleaver Interleaver output input input input input 0 103 175 16 146 1 36 60 163 91 2 155 133 92 63 3 175 11 56 144 4 52 5 111 46 5 130 4 141 12 6 16 70 65 58 7 178 97 118 137 8 141 131 78 25 9 86 80 55 79 10 49 42 5 70 11 129 136 148 33 12 73 50 19 134 13 84 104 153 148 14 142 32 75 66 15 177 75 128 38 16 110 176 32 163 17 8 87 178 118 18 96 109 22 139 19 77 61 156 130 20 139 39 99 72 21 167 107 124 92 22 109 0 4 160 23 2 172 168 23 24 17 23 20 133 25 37 90 115 153 26 146 54 87 128 27 169 160 122 86 28 54 48 9 152 29 134 173 166 106 30 101 27 27 53 31 78 100 155 93 32 135 129 94 61 33 70 14 134 5 34 153 7 38 158 35 6 142 137 172 36 29 20 67 121 37 41 103 161 135 38 143 38 90 44 39 63 126 127 149 40 47 157 43 168 41 124 144 171 0 42 90 21 64 124 43 31 64 162 143 44 152 44 98 27 45 98 79 133 30 46 59 105 34 151 47 133 146 138 114 48 15 49 73 113 49 79 93 154 43 50 164 1 100 138 51 67 84 58 89 52 50 81 103 159 53 128 145 169 17 54 23 18 23 120 55 34 15 117 136 56 154 106 88 102 57 69 91 50 81 58 45 12 13 170 59 9 169 175 176 60 27 63 68 142 61 35 71 39 104 62 156 125 102 21 63 170 37 54 78 64 113 120 37 155 65 127 138 149 8 66 102 17 29 52 67 82 113 150 95 68 149 31 104 62 69 176 130 59 40 70 46 140 3 174 71 13 8 139 6 72 22 25 69 131 73 30 74 110 48 74 163 134 77 18 75 60 115 131 1 76 114 9 42 179 77 11 171 142 34 78 92 46 25 123 79 44 68 158 77 80 157 33 80 26 81 74 116 47 84 82 48 2 35 157 83 132 179 143 85 84 24 52 72 56 85 87 92 151 147 86 140 36 84 67 87 66 78 57 76 88 118 164 8 162 89 123 177 176 10 90 104 24 61 51 91 89 72 46 103 92 136 122 41 140 93 64 118 51 87 94 107 162 10 175 95 14 121 173 115 96 99 16 63 4 97 43 73 107 101 98 115 45 125 69 99 71 53 48 80 100 117 77 11 169 101 12 110 177 75 102 26 30 24 49 103 38 66 30 97 104 147 29 91 154 105 62 76 76 83 106 57 158 109 14 107 131 148 140 2 108 94 111 74 132 109 33 94 114 96 110 151 43 82 16 111 172 83 120 37 112 116 139 1 166 113 10 10 79 109 114 25 56 66 54 115 75 98 119 42 116 144 114 93 28 117 179 117 159 32 118 51 152 36 171 119 120 174 174 119 120 20 47 26 55 121 80 62 112 94 122 160 128 101 65 123 174 85 123 20 124 106 155 44 165 125 1 178 145 3 126 21 26 60 47 127 88 96 157 90 128 137 41 97 117 129 61 82 45 88 130 105 150 33 177 131 5 143 167 11 132 18 58 70 59 133 32 69 152 68 134 158 127 85 73 135 72 86 126 41 136 56 13 40 150 137 125 141 135 111 138 28 35 62 127 139 42 101 108 100 140 161 149 95 110 141 168 108 49 31 142 53 3 31 167 143 7 154 147 13 144 100 51 71 122 145 40 95 113 145 146 145 132 89 71 147 171 135 132 22 148 55 163 6 173 149 3 137 144 116 150 95 28 18 126 151 83 102 105 141 152 162 123 83 29 153 173 112 130 39 154 119 151 2 178 155 126 167 172 57 156 91 59 17 125 157 39 19 164 36 158 150 156 81 19 159 165 119 52 7 160 112 153 7 156 161 122 168 179 107 162 93 55 28 9 163 76 65 160 98 164 138 34 136 74 165 166 6 121 45 166 108 159 14 161 167 121 170 146 112 168 97 57 15 50 169 81 67 106 99 170 148 40 86 24 171 65 89 129 35 172 111 147 12 164 173 4 165 170 64 174 19 22 21 129 175 85 99 116 15 176 159 124 96 60 177 68 88 53 82 178 58 161 0 108 179 0 166 165 105

Table 47 represents a QCB interleaving pattern table when the modulation type is NUC 256 and the code word length is 64K.

TABLE 47 Code 5/15 6/15 7/15 8/15 10/15 11/15 13/15 rate π(j)-th π(j)-th π(j)-th π(j)-th π(j)-th π(j)-th π(j)-th j-th block block of block of block of block of block of block of block of of Group- Group- Group- Group- Group- Group- Group- Group- wise wise wise wise wise wise wise wise Interleaver Interleaver Interleaver Interleaver Interleaver Interleaver Interleaver Interleaver output input input input input input input input 0 39 99 24 85 45 27 59 1 45 100 157 3 31 68 85 2 128 15 0 148 67 35 108 3 84 107 43 161 35 117 128 4 143 54 126 96 159 138 49 5 148 76 172 99 157 83 91 6 2 153 135 154 177 127 163 7 75 174 65 13 2 10 3 8 43 61 32 78 44 60 58 9 50 0 18 160 23 73 16 10 130 36 114 61 73 47 106 11 87 71 42 36 148 115 126 12 137 62 162 21 163 155 74 13 151 137 67 141 118 81 141 14 7 108 104 121 176 170 167 15 71 114 61 115 4 9 35 16 55 65 23 82 14 65 57 17 51 98 11 1 97 66 82 18 133 151 4 59 142 52 30 19 90 19 96 72 37 112 123 20 140 112 163 43 143 150 68 21 149 109 75 135 149 77 95 22 6 152 109 168 179 171 160 23 177 117 58 139 3 161 42 24 37 35 79 46 12 22 62 25 124 93 154 10 32 20 21 26 99 43 3 56 140 39 102 27 83 90 95 67 42 106 131 28 23 154 168 108 167 147 52 29 159 73 73 134 166 90 142 30 0 150 103 111 41 126 157 31 176 165 60 105 126 165 10 32 41 23 84 66 13 23 55 33 121 16 148 89 30 16 79 34 96 91 113 137 144 45 24 35 89 5 40 130 57 113 130 36 30 169 164 104 113 154 73 37 161 175 173 143 147 86 92 38 18 120 143 113 173 173 179 39 172 149 49 11 6 158 2 40 60 26 29 84 52 24 61 41 49 59 156 157 24 71 11 42 134 49 7 32 39 40 104 43 104 56 89 73 64 107 122 44 139 156 132 90 80 136 45 45 166 136 179 38 112 94 140 46 14 110 138 117 104 128 159 47 179 80 53 146 174 163 43 48 62 58 85 53 11 31 148 49 48 55 12 2 151 72 19 50 129 40 117 60 71 33 23 51 105 103 36 93 109 101 111 52 146 159 122 91 162 134 76 53 160 83 66 71 79 80 135 54 16 127 107 114 171 175 169 55 174 111 64 19 127 7 39 56 33 155 28 47 46 61 63 57 54 167 147 4 92 19 77 58 132 11 2 26 38 49 25 59 112 52 90 75 132 111 117 60 145 116 131 109 81 135 75 61 150 142 70 41 120 92 94 62 9 133 144 50 100 130 155 63 77 1 55 153 1 6 5 64 34 2 26 54 53 62 145 65 117 96 15 163 88 74 14 66 92 77 112 31 76 43 26 67 82 86 35 24 60 116 127 68 136 122 128 106 103 133 46 69 165 6 176 42 139 89 138 70 4 131 106 170 99 129 158 71 67 29 59 62 125 8 38 72 36 51 80 80 48 28 64 73 44 21 19 164 93 15 86 74 101 17 6 65 135 34 105 75 81 45 92 128 161 105 118 76 141 126 129 12 77 146 50 77 156 12 174 142 110 84 137 78 3 3 99 167 107 174 175 79 175 168 62 155 121 4 7 80 58 41 82 88 18 32 144 81 47 30 13 8 95 75 84 82 91 37 121 22 69 44 22 83 102 64 41 131 63 118 113 84 32 164 127 158 83 132 54 85 158 78 71 33 111 96 98 86 13 8 139 178 170 169 172 87 178 118 63 145 7 159 9 88 63 113 25 70 16 58 146 89 118 39 151 9 98 18 17 90 100 48 9 51 141 42 27 91 85 140 39 69 61 100 114 92 26 14 159 102 86 141 51 93 167 60 69 140 116 87 139 94 1 82 142 173 172 131 156 95 173 134 52 147 130 157 37 96 38 25 77 83 49 63 147 97 116 33 21 165 25 11 78 98 131 50 119 30 40 48 103 99 107 84 38 126 65 108 115 100 138 28 167 100 87 151 66 101 162 105 178 138 108 79 97 102 8 123 101 171 101 177 168 103 72 145 56 103 5 168 34 104 42 7 87 45 21 26 60 105 115 27 155 159 89 17 83 106 98 34 5 27 75 36 107 107 108 92 91 74 43 102 121 108 24 115 166 97 82 137 48 109 152 147 169 122 146 95 93 110 17 74 146 120 105 122 174 111 171 10 50 16 128 1 33 112 64 68 81 52 17 25 65 113 123 102 20 162 29 21 87 114 94 67 111 132 106 50 99 115 110 63 88 124 34 120 124 116 28 101 165 94 160 153 71 117 147 18 177 133 155 97 136 118 19 66 108 172 175 121 154 119 169 129 47 149 124 0 0 120 61 24 27 86 15 55 150 121 46 4 149 77 28 14 20 122 97 119 115 25 134 46 101 123 106 87 33 68 62 114 112 124 144 42 161 177 119 152 70 125 164 170 72 64 145 91 96 126 5 143 102 174 72 178 170 127 70 121 57 15 10 3 1 128 59 38 86 0 58 30 149 129 53 57 16 125 91 13 80 130 127 95 110 63 74 37 28 131 88 148 97 35 36 103 125 132 31 89 123 34 68 145 53 133 153 81 68 40 150 82 90 134 10 158 100 179 8 125 173 135 73 171 48 20 9 166 6 136 66 32 31 44 54 57 153 137 119 22 14 7 26 76 13 138 126 69 8 55 137 51 29 139 111 53 93 28 56 99 116 140 29 130 130 101 165 144 72 141 155 104 170 150 115 85 88 142 15 161 133 110 114 123 165 143 170 75 44 18 0 162 8 144 57 141 78 119 47 56 143 145 120 9 150 5 27 12 12 146 125 47 118 29 22 53 31 147 80 79 94 76 20 119 119 148 142 162 158 107 168 139 47 149 168 146 76 136 154 78 89 150 11 124 134 112 102 179 164 151 68 157 46 144 123 5 40 152 56 70 83 48 50 64 151 153 52 106 152 81 94 70 81 154 95 31 1 57 66 54 109 155 103 132 37 49 33 110 110 156 27 166 160 92 85 148 44 157 154 128 171 95 59 93 134 158 21 138 136 118 164 172 162 159 78 125 54 17 131 164 36 160 40 44 22 156 51 29 152 161 122 13 17 166 90 69 15 162 93 85 116 23 70 38 100 163 86 88 34 129 138 109 129 164 25 135 125 79 84 143 67 165 163 144 175 37 117 88 133 166 20 173 105 175 178 124 166 167 79 163 45 152 122 160 41 168 35 20 30 87 19 59 56 169 114 46 153 6 96 67 18 170 135 97 10 58 156 41 32 171 109 94 98 127 55 104 120 172 22 139 124 98 78 149 69 173 157 172 74 123 158 98 132 174 12 72 137 39 169 176 161 175 69 160 51 14 129 2 4 176 65 176 120 116 133 167 177 177 74 177 141 169 152 156 176 178 76 178 140 176 136 140 178 179 113 179 145 151 153 142 171

Table 48 represents a QCB interleaving pattern table when the modulation type is NUQ1K and the code word length is 64K.

TABLE 48 Code rate 5/15 7/15 9/15 10/15 11/15 j-th π(j)-th π(j)-th π(j)-th π(j)-th π(j)-th block block of block of block of block of block of of Group- Group- Group- Group- Group- Group- wise wise wise wise wise wise Interleaver Interleaver Interleaver Interleaver Interleaver Interleaver output input input input input input 0 128 117 42 100 33 1 4 61 36 22 73 2 162 46 135 60 90 3 8 179 126 121 107 4 77 24 3 40 99 5 29 161 17 44 94 6 91 142 82 164 53 7 44 133 87 170 151 8 176 11 172 176 124 9 107 6 32 101 8 10 149 121 65 88 12 11 1 44 70 26 117 12 150 103 143 35 21 13 9 76 131 4 58 14 119 22 10 21 158 15 99 63 1 173 77 16 71 136 85 140 72 17 124 151 147 145 59 18 104 33 31 175 123 19 41 8 176 174 2 20 62 123 66 81 125 21 5 60 47 28 157 22 118 105 97 72 50 23 50 175 128 112 62 24 174 18 8 132 109 25 54 160 9 106 75 26 111 138 146 42 42 27 40 147 73 56 146 28 156 10 162 151 118 29 92 0 164 147 153 30 46 125 57 82 85 31 11 57 64 49 10 32 17 49 139 91 131 33 52 75 91 64 70 34 47 21 5 179 32 35 97 154 110 89 41 36 179 140 150 160 24 37 24 150 83 52 143 38 153 9 18 139 113 39 145 169 27 17 1 40 129 124 48 97 93 41 2 55 45 63 162 42 12 48 133 116 20 43 88 173 132 131 35 44 101 23 111 154 74 45 139 157 124 71 45 46 114 97 89 109 149 47 69 129 78 96 161 48 96 30 177 135 173 49 32 7 19 146 4 50 134 122 46 55 28 51 55 54 50 38 23 52 167 99 102 166 127 53 132 74 103 117 148 54 123 19 122 65 34 55 136 153 4 127 61 56 112 94 74 120 96 57 102 128 161 129 144 58 159 15 175 15 171 59 31 170 34 136 140 60 87 87 60 74 119 61 141 59 58 23 16 62 15 51 136 98 126 63 61 80 100 43 39 64 84 111 115 123 40 65 98 64 118 130 57 66 37 137 81 69 165 67 63 146 75 99 106 68 20 13 28 143 172 69 85 2 21 161 139 70 53 83 40 46 81 71 7 62 61 51 47 72 39 45 140 94 164 73 117 176 138 61 92 74 170 108 113 83 63 75 138 71 112 67 105 76 116 91 157 156 108 77 126 131 151 33 170 78 161 34 23 144 3 79 120 168 30 148 135 80 57 82 69 163 101 81 13 56 41 47 121 82 76 102 94 92 68 83 6 72 96 2 6 84 121 26 7 122 111 85 155 155 109 24 65 86 175 92 152 86 147 87 38 132 149 75 150 88 158 31 33 108 122 89 35 166 179 152 7 90 86 119 71 14 84 91 78 36 43 77 46 92 10 101 92 7 22 93 103 178 105 10 103 94 166 113 12 29 86 95 95 67 13 19 169 96 125 98 154 104 134 97 172 152 159 128 44 98 67 14 178 142 175 99 30 5 24 1 167 100 177 118 44 79 89 101 73 41 49 107 128 102 151 104 107 162 27 103 169 177 98 0 31 104 163 114 16 118 56 105 23 70 2 66 43 106 108 96 76 54 102 107 43 134 155 153 156 108 81 32 35 141 160 109 157 162 168 9 141 110 58 84 62 85 67 111 105 40 56 37 9 112 65 100 129 32 110 113 26 174 141 114 159 114 122 110 116 53 133 115 135 158 123 134 78 116 146 93 160 41 154 117 72 149 77 158 176 118 142 27 25 178 174 119 34 4 170 138 5 120 133 86 54 76 82 121 0 38 39 50 11 122 148 53 90 78 25 123 89 77 95 84 80 124 168 115 121 172 130 125 60 159 11 48 163 126 109 143 72 133 88 127 83 130 153 168 36 128 18 35 169 125 166 129 27 163 167 13 137 130 131 89 51 169 104 131 70 58 67 25 48 132 56 106 104 16 129 133 48 73 134 8 87 134 64 20 0 124 95 135 93 66 117 159 55 136 68 90 79 167 49 137 127 127 80 58 145 138 21 16 26 5 178 139 75 3 29 11 0 140 110 85 37 68 98 141 80 37 55 95 64 142 14 107 99 27 54 143 49 172 142 110 100 144 82 116 108 93 37 145 143 156 114 62 79 146 115 95 86 102 69 147 178 144 88 137 38 148 154 17 166 126 177 149 100 165 163 150 136 150 59 81 59 87 114 151 74 43 63 105 17 152 152 50 101 113 52 153 51 78 93 30 19 154 137 109 119 119 30 155 140 68 15 6 97 156 36 135 144 103 51 157 42 126 145 57 168 158 19 29 165 31 132 159 25 167 22 149 138 160 94 120 52 80 83 161 45 39 53 70 76 162 164 47 130 45 13 163 16 171 137 165 18 164 113 112 125 111 115 165 79 69 6 73 71 166 22 141 158 36 91 167 28 145 84 157 179 168 66 28 20 171 112 169 106 1 174 3 155 170 130 88 38 20 15 171 171 42 68 18 14 172 147 52 127 90 26 173 90 79 106 12 60 174 144 25 14 59 29 175 165 65 120 39 116 176 3 139 148 115 66 177 173 148 156 34 120 178 160 12 171 177 142 179 33 164 173 155 152

Both apparatus and method inventions are mentioned in this specification and descriptions of both of the apparatus and method inventions may be complementarily applicable to each other.

Various embodiments have been described in the best mode for carrying out the invention.

The present invention is available in a series of broadcast signal provision fields.

It will be apparent to those skilled in the art that various modifications and variations can be made in the present invention without departing from the spirit or scope of the inventions. Thus, it is intended that the present invention covers the modifications and variations of this invention provided they come within the scope of the appended claims and their equivalents. 

What is claimed: 1-20. (canceled)
 21. A method for transmitting broadcast signals by an apparatus for transmitting broadcast signals, the method comprising: encoding service data; bit interleaving the encoded service data, wherein the bit interleaving includes: first interleaving the encoded service data, wherein the first interleaving interleaves parity bits of the encoded service data; second interleaving a number of blocks in the first interleaved service data by using an interleaving pattern for a combination of a 16 Quadrature Amplitude Modulation (QAM) modulation type, a 6/15 code rate, and a 16200 code word length of the encoded service data, and wherein each block has 360 bits, wherein the interleaving pattern is represented as 6/15 π(j)-th j-th block of block of Group- Group- wise wise Interleaver Interleaver output input 0 12 1 13 2 15 3 30 4 27 5 25 6 11 7 34 8 9 9 4 10 31 11 22 12 6 13 32 14 7 15 21 16 17 17 3 18 1 19 26 20 10 21 33 22 19 23 2 24 18 25 5 26 28 27 35 28 8 29 16 30 29 31 23 32 14 33 0 34 20 35 24 36 36 37 37 38 38 39 39 40 40 41 41 42 42 43 43 44 44

wherein j-th block of the number of blocks is interleaved by a block unit according to the interleaving pattern; third interleaving the second interleaved service data, wherein the third interleaving performs: row-wise writing bits of each block in the second interleaved service data, wherein a number of rows is based on the modulation type, and column-wise reading out 1 bit of the written each block, wherein a number of columns corresponds to 360; building at least one signal frame including the bit interleaved service data; modulating data in the built at least one signal frame by an Orthogonal Frequency Division Multiplex (OFDM) scheme; and transmitting the broadcast signals having the modulated data.
 22. The method of claim 21, the method further includes: mapping the bit interleaved service data to symbols based on the 16 QAM.
 23. The method of claim 22, wherein the service data is encoded according to the code rate.
 24. The method of claim 23, the method further includes: time interleaving the bit interleaved service data based on a skip operation.
 25. An apparatus for transmitting broadcast signals, the apparatus comprising: an encoder to encode service data; a bit interleaver to bit interleave the encoded service data, wherein the bit interleaver further performs: first interleaving the encoded service data, wherein the first interleaving interleaves parity bits of the encoded service data; second interleaving a number of blocks in the first interleaved service data by using an interleaving pattern for a combination of a 16 Quadrature Amplitude Modulation (QAM) modulation type, a 6/15 code rate, and a 16200 code word length of the encoded service data, and wherein each block has 360 bits, wherein the interleaving pattern is represented as 6/15 π(j)-th j-th block of block of Group- Group- wise wise Interleaver Interleaver output input 0 12 1 13 2 15 3 30 4 27 5 25 6 11 7 34 8 9 9 4 10 31 11 22 12 6 13 32 14 7 15 21 16 17 17 3 18 1 19 26 20 10 21 33 22 19 23 2 24 18 25 5 26 28 27 35 28 8 29 16 30 29 31 23 32 14 33 0 34 20 35 24 36 36 37 37 38 38 39 39 40 40 41 41 42 42 43 43 44 44

wherein j-th block of the number of blocks is interleaved by a block unit according to the interleaving pattern; third interleaving the second interleaved service data, wherein the third interleaving performs: row-wise writing bits of each block in the second interleaved service data, wherein a number of rows is based on the modulation type, and column-wise reading out 1 bit of the written each block, wherein a number of columns corresponds to 360; a frame builder to build at least one signal frame including the bit interleaved service data; an modulator to modulating data in the built at least one signal frame by an Orthogonal Frequency Division Multiplex (OFDM) scheme; and a transmitter to transmit the broadcast signals having the modulated data.
 26. The apparatus of claim 25, the apparatus further includes: a symbol mapper to map the bit interleaved service data to symbols based on 16 QAM.
 27. The apparatus of claim 26, wherein the service data is encoded according to the code rate.
 28. The apparatus of claim 27, apparatus further includes: a time interleaver to time interleave the bit interleaved service data service data based on a skip operation.
 29. A method for receiving broadcast signals by an apparatus for receiving broadcast signals, the method comprising: receiving the broadcast signals; demodulating the received broadcast signals by an Orthogonal Frequency Division Multiplex (OFDM) scheme; parsing at least one signal frame from the demodulated broadcast signals; bit deinterleaving data in the parsed at least one signal frame, wherein the bit interleaving is a reverse process of a bit interleaving, wherein the bit interleaving includes: first interleaving the encoded service data, wherein the first interleaving interleaves parity bits of the encoded service data; second interleaving a number of blocks in the first interleaved service data by using an interleaving pattern for a combination of a 16 Quadrature Amplitude Modulation (QAM) modulation type, a 6/15 code rate, and a 16200 code word length of the encoded service data, and wherein each block has 360 bits, wherein the interleaving pattern is represented as 6/15 π(j)-th j-th block of block of Group- Group- wise wise Interleaver Interleaver output input 0 12 1 13 2 15 3 30 4 27 5 25 6 11 7 34 8 9 9 4 10 31 11 22 12 6 13 32 14 7 15 21 16 17 17 3 18 1 19 26 20 10 21 33 22 19 23 2 24 18 25 5 26 28 27 35 28 8 29 16 30 29 31 23 32 14 33 0 34 20 35 24 36 36 37 37 38 38 39 39 40 40 41 41 42 42 43 43 44 44

wherein j-th block of the number of blocks is interleaved by a block unit according to the interleaving pattern; third interleaving the second interleaved service data, wherein the third interleaving performs: row-wise writing bits of each block in the second interleaved service data, wherein a number of rows is based on the modulation type, and column-wise reading out 1 bit of the written each block, wherein a number of columns corresponds to 360; decoding the bit deinterleaved data.
 30. The method of claim 29, the method further includes: demapping data in the parsed signal frame based on the 16 QAM.
 31. An apparatus for receiving broadcast signals, the apparatus comprising: a receiver to receive the broadcast signals; a demodulator to demodulate the received broadcast signals by an Orthogonal Frequency Division Multiplex (OFDM) scheme; a frame parser to parse at least one signal frame from the demodulated broadcast signals; a bit deinterleaver to bit deinterleave data in the parsed signal frame, wherein the bit deinterleaving is a reverse process of a bit interleaving, wherein the bit interleaving includes: first interleaving the encoded service data, wherein the first interleaving interleaves parity bits of the encoded service data; second interleaving a number of blocks in the first interleaved service data by using an interleaving pattern for a combination of a 16 Quadrature Amplitude Modulation (QAM) modulation type, a 6/15 code rate, and a 16200 code word length of the encoded service data, and wherein each block has 360 bits, wherein the interleaving pattern is represented as 6/15 π(j)-th j-th block of block of Group- Group- wise wise Interleaver Interleaver output input 0 12 1 13 2 15 3 30 4 27 5 25 6 11 7 34 8 9 9 4 10 31 11 22 12 6 13 32 14 7 15 21 16 17 17 3 18 1 19 26 20 10 21 33 22 19 23 2 24 18 25 5 26 28 27 35 28 8 29 16 30 29 31 23 32 14 33 0 34 20 35 24 36 36 37 37 38 38 39 39 40 40 41 41 42 42 43 43 44 44

wherein j-th block of the number of blocks is interleaved by a block unit according to the interleaving pattern; third interleaving the second interleaved service data, wherein the third interleaving performs: row-wise writing bits of each block in the second interleaved service data, wherein a number of rows is based on the modulation type, and column-wise reading out 1 bit of the written each block, wherein a number of columns corresponds to 360; a decoder to decode the bit deinterleaved data.
 32. The apparatus of claim 31, the apparatus further includes: a demapper to demap data in the parsed signal frame based on the 16 QAM. 